DIYstompboxes.com

Jordan Bosstone
Ok, So this one is a little out of the ordinary.
It uses NPN and PNP transistors.

(http://www.basicaudio.net/Boss-Tone-tech.jpg)

R1 is a voltage divider which turns down the signal into the circuit.
C1 forms a high pass filter with R5
R3 and R4 form a feedback resistance and filter along with C2
which also sets the bias for the base of Q1 withe the help
of R5 (R3+R4 and R5, voltage divider).
C3 filters out some high end and possible oscillation.
Q1 (NPN) is feed directly into Q2 with no coupling cap. ("direct coupled"?
Q2 is PNP is set up with the emmiter to 9v through R2.
So does this make Q2 a buffer of sorts? This is where I bog down...
Anyone care to tackle this?

Also. When the Attack pot is lowered oscillation is somewhat common
in the builds I've done. Why is this?

John

Not much to add, but my bosstone has oscillation when the gain is over 90% up, there isn't much more gain in the last 10% so I am not missing much.  I had to add a 10uf cap from positive to ground in order to cut down on oscillation and to stop the most annoying thing, it always picked up radio stations, even when boxed up.  This probably wasn't what you were looking for, but there it is.

Yes, I remember thinking why did they use a PNP on the end? It's an emitter follower, so an NPN could have been used.

I think there is more going on in here.
Even without the diodes this thing is pretty fuzzy.
More than just one low hfe transistor and an an emitter follower
can do. Hmmm....

Just for my own clarification and not really adding something insightful or important here but, with those diodes on the end the output is limited to + - Diode drop right? So no matter how gigantic a signal should be coming out it will be between like .7 and -.7 or .3(.24) if they are germanium?

Hi

QuoteIt's an emitter follower, so an NPN could have been used.

The emitter of Q2 takes 50% (18k/36k) of the output voltage of Q1's collector.  That makes Q2 an active load on Q1 because the laws of physics say that the base of a PNP transistor is about 0.65 V lower than its collector.  This has some really cool effects in this circuit.  The gain is high and for small signals (0.02 V p-p), the slew rate at the output of Q2 is very high.  The results would sound very harsh except for 2 things.  Firstly, Q1 is *barely* conducting, and the base voltage is only 0.6V (or slightly less - 0.58 V ?).  This means that "upward" swings (voltage increases) on the base make Q1 conduct better.  Downward swing causes cut off.  This gives the output (pre clipping diodes) an asymetric appearance (like a FuzzFace).  A sine wave results in a mark-space ratio of about 2:1 for small signals.  A bit like a tube-amp. I was surprised that the clipping diodes do not simply "cut" the signal at + and - 0.65 V.  They leave some overshoot, especially on the low side (maybe because the output impedance is different on the + and - sides?).

The circuit seems quite insensitive to the values of many components.  22k replaces 18k, 1 Meg and a 22uF bypass cap can replace the old-fashioned 2x560k split with a small cap.

The results are quite sensitive to the amplitude of the signal.  All in all, an interesting and attack-responsive circuit.

cheers

PS Here's a LTspice approximation of the Bosstone:
R1 N002 N006 1Meg
R2 N006 0 220k
R3 N002 N004 22k
R4 N003 0 100k
Q1 N004 N006 0 0 2N2222
C1 N006 N005 0.022µF
C2 N004 N002 47pF
C3 N002 N003 0.022µF
V1 N005 0 SINE(0 0.1 880) AC 0.1
V2 N001 0 9 Rser=10
D1 N003 0 1N4148
D2 0 N003 1N4148
R6 N002 N001 22k
Q3 0 N004 N002 0 2N3906
C4 N001 0 22µ V=20 Irms=0 Rser=0.225 Lser=0 mfg="KEMET" pn="T495D226M020AS" type="Tantalum"
.model D D
.lib C:\PROGRA~1\LTspice\lib\cmp\standard.dio
.model NPN NPN
.model PNP PNP
.lib C:\PROGRA~1\LTspice\lib\cmp\standard.bjt
* Jordan Bosstone
;op
.tran 0 10ms 0
.backanno
.end

Mine hasn't oscillation but with the pot at max, the sound become really odd . I must keep the pot at 80 per cent , no more, to hear the good fuzz sound.

Brett

"The circuit seems quite insensitive to the values of many components.  22k replaces 18k..." 

I'm assuming that you're talking about R2 from the schematic John posted...  I did find that lowering slightly (16k or 15K)  helped tame some of the oscillation problems people report (while not drastically altering the tone of the pedal).

Jesse

Thanks for the insight brett.

I haven't had any oscillation with the attack full up but it's a somewhat common problem with
oscillation when the attack is turned all the way down. Voodoo labs fixed this with a 1k resistor between
the bottom of the attack pot and ground, setting  a minimum atttack amount before osc.
Why does this happen? Something to do with shunting R5 to ground affecting bias at the base?
Or maybe something to do with the R3/R4/C2 filter?

I have an original one and it doesn't have that problem, maybe there is something wrong in the schematics going around.

Al

Hi

QuoteI'm assuming that you're talking about R2 from the schematic John posted...  I did find that lowering slightly (16k or 15K)  helped tame some of the oscillation problems people report (while not drastically altering the tone of the pedal).

Yes, R2 and R6 can be changed to 15k or 22k if you don't have 18k resistors.  (and who has 50pF caps?  47pF is the obvious replacement, but it might be worth experimenting with higher values, too).

I'm no expert, but I wonder if the oscillation is because Q1 is barely conducting and because new transistor have high hFE than older ones (making the Beta of Q1 more like 300 or 400 rather than a vintage vaue of 150 or 200).  My thinking is this:  Even without an emitter resistor, the effect of being barely turned on creates an input resistance that is extremely high (Rin = Beta*0.025/Ie, where I is the emitter current).  If beta is 400 and Ie=1uA, then Rin = 10 megohms (!).  That is perfect for picking up some of the output signal.  Therefore, if I had oscillation issues, I'd try lower hFE transistors (TIPs or BD139s) and be careful to separate the input and output traces on the layout (and perferably not use vero).
cheers

My Bosstone clone performs nicely but starts to produce weird oscillations as the battery wanes.  Not sure why, but it does so reliably.

Brett,

Don't sell yourself short!  You have made many posts concerning the Bosstone.  Not to hijack the thread and make it about oscillation, but... my current breadboarded version uses low hfe transistors and still is prone to oscillation when the attack pot is turned to a particular spot and I hit a particular note. It's pretty specific.   

Mark's post will make me run and measure the battery's voltage-- I have a suspicion that it may be low, and that by lowering the resistance of R2, it helped remedy the situation just enough... 

There's a lot of resonance in the circuit... almost has a twin-t wah filter sound.  There seem to be similarities in the Q1 stages of both circuits (I'm still learning, though).  My original build when going through lower settings of the attack pot sounded like a %^&*ed wah a bit before becoming a fuzz monster.  In that build I used an NTE123A in Q1, and a 2N4125 in Q2.  All of the resistor values were identical to the schematic, as were the cap values (except for a  47pF).  Sounds awesome-- I regret getting rid of it, but will hopefully have another one up and running soon.

Jesse

Great little fuzz.  I have a 3-position switch in mine that selects between an increase in input cap value (for that quasi-octave-down sound), stock, and a cap in parallel with the clipping diodes for a "rounder" sound.

I built one of these for a friend a few years ago and he never had a problem with it.
Well, except for minor 'tweaks' he was constantly pestering me for...  :icon_mad:

There is another version posted at the free information society:
http://www.freeinfosociety.com/electronics/schemview.php?id=933

From ToneFrenzy's site (http://www.tonefrenzy.com/effects/jordan_boss_tone.html) I gather there were two versions of this pedal.
One would plug directly into your guitar like one of Dan Armstrong's 'jack warts' and a pedal version a few years later.

I wonder which schematic is which?  :icon_question:

Hi
is my memory playing up, or do I recall the two circuits being called the "Nashville" and the "Dallas" versions (or some similar geographic references?).

I haven't had a problem with oscillations, but I made 2 chages.  Firstly, I used a 1 M resistor to bias Q1.  This would turn is on a bit more and make it a bit less buzzy.  This would reduce the input resistance which might be the source of the feedback (see my post above).  I also put a 10 ohm resistor on the emitter of Q1.  This may have knocked the gain down a bit (and lifted the base voltage just a tiny bit further).

My guess is that the 2 x 560k resistors could be replaced with 2 x 470k without losing much (any?) Bosstone magic, while improving oscillation issues and probably making the circuit less sensitive to the choice of BJT for Q1.

cheers

There are at least two versions.
Nashville, CA and one other than I can't think of now...

(http://www.aronnelson.com/gallery/main.php?g2_view=core.DownloadItem&g2_itemId=18408&g2_serialNumber=2)

Here's a "factory" schematic

(http://www.aronnelson.com/gallery/main.php?g2_view=core.DownloadItem&g2_itemId=8647&g2_serialNumber=2)

Quote from: John Lyons on August 09, 2009, 10:42:55 AM
Thanks for the insight brett.

I haven't had any oscillation with the attack full up but it's a somewhat common problem with
oscillation when the attack is turned all the way down. Voodoo labs fixed this with a 1k resistor between
the bottom of the attack pot and ground, setting  a minimum atttack amount before osc.
Why does this happen? Something to do with shunting R5 to ground affecting bias at the base?
Or maybe something to do with the R3/R4/C2 filter?

I think it might have something to do with phase shift.

I think it might have something to do with phase shift.
Any further theory on this?

Just speculating on my part, but for fun follow this path and ignore the other components:

NPN collector, up the 18k resistor, stop.  Over to one side is a .022 cap to ground.  Turn left and head down the 560k resistor, stop.  Hey, another .02 cap to ground!  Continue heading toward the NPN's base along the next 560k resistor, then stop at the base.  With the input control grounded, why look!  There's another .02 cap to ground!  We got ourselves a phase-shift oscillator!

Now add back in the other parts.  The PNP follower causes the 18k resistor to have the same signal appearing at both ends, thus less current can flow through it and it "appears" larger than it really is.  This would be good for making an oscillator, as it would make this resistor closer in value to the two 560k resistors that make up the other parts.  However, there's a wrinkle here because the PNP emitter is also where the first cap was located, and on the other end of that cap are a couple diodes and a volume pot.  For small signals coming through the cap, ground is 100k away, not exactly a short circuit.  But for larger signals, the diodes conduct and there ya go.  Then there's still that pesky PNP emitter.  To a small signal it looks like a low impedance input which would tend to stop any signal from proceeding any farther.  This is where my "theory" runs out of steam.  To make a "good" oscillator out of this circuit, I'd probably want to put some series resistance between the emitter of the PNP and the .022 output cap.

This may account for why some people have oscillations and others do not.  Different layouts and different part tolerances might make up the difference.

Interesting, thanks for the rundown. I'll have to think on that...

John

In simulation, I got oscillations, and taking C2 out seemed to stop it. The voltage at that junction swings up and down quite dramatically without the cap; with the cap, the voltage only goes up and down half a volt or so, which holds the base of Q1 higher than it "should" be, so the gain goes up dramatically. However, with the cap oscillation can take place for the reasons earthtonesaudio mentioned above.

Next, if we remove Q2 for a second, then the north end of C4 would bounce up and down between 4.5V or so and 8.5V. Put Q2 back in and we find that the voltage drop across R6 (refer to the schem at the start of the thread) is constrained to be about 0.6V or so. This means that, as Q1 conducts heavily, the collector falls to almost 0V. That collector is connected directly to Q2's base, so Q2's emitter drops to about 0.7V. On the up-swing, Q1's collector shoots up to about 8.5V. This happens with signals greater than about 30mV. With smaller signals, Q2's base and emitter continue to just track up and down 0.6V apart in time with the voltage at Q1's collector and don't really have any effect apart from that.

C4 is not simply there to de-couple the output. If you take C4 out, the circuit simply refuses to output any signal at all. If you simply replace C4 with a wire, you now have 9V ->18k (R2 in the original schem) -> diode ->GND, so the emitter of Q2 and the "north" ends of C3, R3 and R6 are all held firmly at about 0.6V, so absolutely nothing happens. Putting C4 in there means that the rest of the circuit can swing up and down to its heart's content. Take the diodes out for a second, so we can think about what happens at the right hand end of C4. As the voltage at Q2's emitter swings high, C4 charges up. Then the Q2E voltage takes a nose-dive. That charge has to go somewhere, so the voltage at the right hand end of C4 goes below zero. As stated, the 100k pot is not exactly a short-circuit, so we get a weird, but quite smooth waveform at the output. Put the diodes back in and clipping occurs any time the output exceeds +0.6V or drops below -0.6V. So, outside those bounds we get hard limiting of the output signal at + or - 0.6V as the diodes conduct heavily. Inside those bounds, the current increasingly has to flow through the 100k pot to ground. This is where the very weird waveforms come from with this circuit.

Hi
Yes. And that further explains why I haven't had the oscillations.  I replaced that cheapo and outdated 2x560k and a small cap with a more modern 1M and a 22uF decoupling cap.

Yep, the waveforms can be really weird.  I *think* much of that is because the output impedance is different on the + and - sides of the signal. 
cheers

Just putting the schem on this page for easier reference.

(http://www.basicaudio.net/Boss-Tone-tech.jpg)

Thanks for the simulation R O!
I'll have to digest this.

john

Nashville version:

(http://i3.photobucket.com/albums/y57/c-martin/fuzzes/close%20shots/jordan-bosstone-nash-compside.jpg)

(http://i3.photobucket.com/albums/y57/c-martin/fuzzes/close%20shots/jordan-bosstone-nash-trace.jpg)

Alhambra version:

(http://i3.photobucket.com/albums/y57/c-martin/fuzzes/close%20shots/jordan-bosstone-ca-compside.jpg)

(http://i3.photobucket.com/albums/y57/c-martin/fuzzes/close%20shots/jordan-bosstone-ca-circuit.jpg)

Hey now!
I dig the ceramic cap "pancake stack" thing they have going.
Perhaps that's to induce some parasitic capacitance within the stack to cut the feedback?
Or just to make room?

Hmmm....

It also appears that the number of capacitors gives away the variety:
3 caps = Nashville
2 caps = Alhambra

Thanks for the shots Chris!

If my soldering looked like that, I'd be ashamed of myself...

OK, done a bit more simulation and it appears that placing a 22k resistor between the north end of C2 and the junction of R3/R4 damps out the oscillations without affecting the response unduly.

I also simulated a guitar pickup at the input complete with Vol and Tone, rather than a simple ideal AC signal source, because it occured to me that the resonant circuit that they make up might be the culprit if this was the first or the only pedal in the chain. Didn't make any difference one way or the other. Without the damping resistor it oscillated, with it, it didn't.

I'd be interested to see if someone who has a "howler" could try this damping resistor thing and see if it actually works.

Hi

Quoteit appears that placing a 22k resistor between the north end of C2 and the junction of R3/R4 damps out the oscillations without affecting the response unduly

Although that will reduce or eliminate the oscillations, it also minimizes the power supply filtering.  C2 is an old-school troublemaker that does nothing tonally. Ditch it and connect a 10/22/47 uF cap from V+ to ground.
cheers

Quote from: brett on August 20, 2009, 08:03:21 PM
Hi

Quoteit appears that placing a 22k resistor between the north end of C2 and the junction of R3/R4 damps out the oscillations without affecting the response unduly

Although that will reduce or eliminate the oscillations, it also minimizes the power supply filtering.  C2 is an old-school troublemaker that does nothing tonally. Ditch it and connect a 10/22/47 uF cap from V+ to ground.
cheers

C2 cuts negative feedback from the collector to the base, which increases gain above the 3dB rolloff created by R4/C2 (12Hz) and also increases the input impedance (though not significantly because R5 dominates).  It also filters power supply noise.

Adding the 22k in series with C2 make a voltage divider in conjunction with R4 so that the gain is (neglecting active load details) R4/22k.

The trouble with the 1M/22µF mod is that the gain drops unacceptably. C2 might not affect the tone, but it sure affects the gain in spades.

Adding the 22k in series with C2 does, indeed, reduce the gain but, as I said earlier, "not unduly". I've done some more tweaking and 4k7 might well be enough to damp things out and not affect the gain at Q1's collector more than a hair.

C2 is a voltage divider to AC signals in conjunction with R4 in any case, so it's just a matter of the degree to which the extra resistor affects things. Looking at the numbers, the %age gain change is small in the range 82Hz - 1.3kHz (low E fundamental to 24th fret on high E fundamental). Above that frequency, C2's reactance becomes quite small, so the new resistor begins to dominate. At 82 Hz, C2's reactance is 88k. At 400Hz, 18k, at 1kHz, 7.2k and 2kHz it's 3.6k. It's hard to say what effect this limit on the higher harmonic content's gain will have. Let's face it, the output waveform is totally FUBAR'd in comparison with the input waveform in any case :) so who can say what's normal and what isn't? With that 4k7 in there, a mere +/-1mV input signal @ 800Hz gives +/-625mV at Q1's collector. That's an insane gain of 625! Short the resistor out and you get a gain of 640. Take C2 out altogether and your gain drops to 120.

As to power supply filtering? This thing draws only 350µA or so. I'd run it off a battery, I think. TBH, a standard 9V battery will last months of "normal" use at that current draw. If you really wanted to run it off a PSU, why not a 10µF || 100nF (or similar) from +9V to GND as you suggest, Brett? I agree totally.

But why hamstring this insane pedal by overly limiting its gain (taking out C2) vs giving it just a little more damping than was originally designed into it and otherwise letting it do its funky thang?

I'm on my 3rd Bosstone build and my results are somewhat confusing.

The 1st 2 were great distortion pedals but not fuzzy at all. This 3rd one is fuzzy but has no volume.  I swapped transistors, checked all my connections, made sure my soldering was good, checked my pots... even went as far as to build another board with all new parts... the problem remains.

I've tried a few different trannys for Q1 but have pretty much stuck to the 3906 for Q2.

I don't even know if I'm looking for an answer here or just maybe someone else who has built this thing wth less than fuzzy results.

Yes, the bosstone is more of a distortion/fuzz.
If you used all new parts and a new board then
it's got to be the schematic or something wrong
that you are doing when you build it.

There aren't any other variables unless the parts
are bad somehow.

john

I was looking the factory schem, r2:15 and c3:50uf ???

IMHO it works better with lower gain transistors as brett noted, no more than 150.
And can be fine tune by tweaking the 150k resistor. Put a 250k trim to find the sweet spot.
I went even further, I used Ge transistors :)

mac

i've been doing some interesting things with the bosstone lately as well.  messing around with Ge trannies and using a variation of mac's diode biasing scheme.  i'll post a schematic when i get home later along with some waveforms that i got.  i found a VERY odd quirk within this circuit as well.

this is the schematic of what i have been messing with.  a silicon booster stage into a Ge bosstone sans diode clippers.  i tied a pot to the C and E of Q2 in order to be able to vary the shape of the wave and the phase of the output.  some cool sounds can be had this way.  you can see that on Q3 i used mac's diode biasing method to get the circuit to conduct properly.  the odd thing that i found was that when the collector of Q3 was lifted, i actually got more output out of the collector.  matter of fact it sounds really nice.  with the collector grounded, you get a waspy sixties fuzzrite kind of sound, lift the collector and you get a nice full fuzz.  the signal off of the emitter is less distorted, but sounds good none the less.  with the phase adjustment and the lift switch you can easily get 5 or 6 very different tones.  below are the wave forms.  can you guys tell me what is happening when C of Q3 is lifted and why there is a much stronger signal off of the collector?  i'll post some of the waveforms so you can see what i am talking about.

(http://i75.photobucket.com/albums/i293/rnfr/BOSSTHING.png)

COLLECTOR GROUNDED, COLLECTOR OUTPUT.
(http://i75.photobucket.com/albums/i293/rnfr/GRCOL.jpg)
COLLECTOR LIFTED, COLLECTOR OUTPUT.
(http://i75.photobucket.com/albums/i293/rnfr/LIFTCOL.jpg)
COOLECTOR LIFTED, EMITTER OUTPUT.
(http://i75.photobucket.com/albums/i293/rnfr/LIFTEMIT.jpg)
COLLECTOR GROUNDED, EMITTER OUTPUT.
(http://i75.photobucket.com/albums/i293/rnfr/GREMIT.jpg)

sorry for the crooked waves, i just got the scope and it needs adjusting.

Quotesorry for the crooked waves, i just got the scope and it needs adjusting.

at least you have  a scope!!! :)

mac

yeah, i got it for 40 bucks cheap!  it's a lot of fun.  these were actually the first waveforms i got out of it.  any idea what's going on here?  is the third trannie just acting as a clipper when lifted??  maybe the 18K resistor is reducing the gain? that batman looking curve actually sounds really nice.  very harmonic percolator-ish.  the others aren't bad either.  the emitter outputs sound nice and overdrivey- very coll with clippers added too.

it always seemed to me that the second transistor, or third in your schem, is acting like a clipping diode, and that the 18k to vcc makes it conduct a little.
a kind of bazz fuss + output clipping diodes.
but unlike the bazz fuss the base-emiter path is not between C and B but in series with another 18k. there is a post about something similar, a very simple circuit like a BF but it has a 10k and a diode and the output taken from the junction of both. it produces an octave up fx.  sorry i do not remember the link, i have to search for it.

mac

Okay, awesome thread. I'm about to start building a Bosstone. I'd like to order all the parts at once. I'd like to order the part(s) needed to rid of any oscillations, should they arise (as they seem to be common).

I was going to use this layout:

(http://aronnelson.com/gallery/main.php?g2_view=core.DownloadItem&g2_itemId=2661&g2_serialNumber=2)

Any word on what I would need to add to this circuit to stop oscillations? Yes, I read the thread, and there is a ton of tech info that is a bit over my head. Is there a definitive answer on how to do this?

Much appreciated!

Thanks!

Or perhaps I should use this layout? Whats with the 2k7 resistor? and the 18k wired to Q2?

(http://aronnelson.com/gallery/main.php?g2_view=core.DownloadItem&g2_itemId=40050&g2_serialNumber=2)

I'd like to order the part(s) needed to rid of any oscillations, should they arise (as they seem to be common).

Use low gain transistors. 200ish hfe or lower

Right, 2n2222 and 2n3906 are under 200 hfe, yeah?

Also, will either of those layouts work? They seem to have differences. Like the 2.7k resistor, etc.

Thanks!

Hi
modern 2N3906s are way over 200.  Even 2N2222s are too high for older circuits.
Try BD139s or other medium power (2 to 10W) devices instead of old small-signal transistors.  hFE around 150.  TIPs, MJEs, etc are also good.
cheers

QuoteJust speculating on my part, but for fun follow this path and ignore the other components:

NPN collector, up the 18k resistor, stop.  Over to one side is a .022 cap to ground.  Turn left and head down the 560k resistor, stop.  Hey, another .02 cap to ground!  Continue heading toward the NPN's base along the next 560k resistor, then stop at the base.  With the input control grounded, why look!  There's another .02 cap to ground!  We got ourselves a phase-shift oscillator!

IIRC gain has to be above 29 for stable oscillations. Can you estimate maximun hfe to be below that limit?

mac

Quote from: Sir H C on August 20, 2009, 11:21:31 AM
Nashville version:

(http://i3.photobucket.com/albums/y57/c-martin/fuzzes/close%20shots/jordan-bosstone-nash-compside.jpg)

My apologies for the thread necromancy, but is that a 2n3565?  I hadn't seen a version that used that, maybe I just need to read up on this circuit some more.

most likely.
from my notes the Nashville used a 2N3565 and a M924/95232, though there may have been other combos used(?).
the gains of 2N3565s can vary quite a bit though, mine go from hfe 150 to 600, so I'd make sure to use a low gain one if you use one.

It's alive! It's aliiiive!

So Ive been trying to test the versatility of this old favorite of mine by playing around with different kinds of gain and tone control. One thing that may be interesting to some people: I had oscillations in certain combinations of gain, tone and starve control (like in the Trombetta/Rockett WTF, highly recommended mod for this one, sounds fantastic and makes it much more versatile, going from a farty mess to brutal imploding duck-and-swell fuzz to rather nice distortion). Placing a 3n9 cap across the clipping diodes killed that dead without cutting an awful lot of highs.

But there is something that I just now realized I don't understand: What the actual funk does C3 do? If Q2 were an NPN and upside down, then C3 would be between Base and Collector and make perfect sense as a treble cut on Q2 (as seems to be claimed or insinuated in all threads on the topic I found here and elsewhere). But it's not. As it is C3 is between Base and Emitter of a common collector stage, and since V_E = V_B there is no voltage across C3 since both its ends move in unison. So all it does is bypass R6 from the perspective of Q1. This does indeed cause a bit of a high cut above 176kHz. Have I gone completely nuts now, or is this the most woefully ineffective way of filtering out radio frequencies? Wouldn't placing this cap from Base or Q1 to ground or a smaller cap between Emitter and Base of Q1 make a lot more sense? There also seems to be some confusion about the value of C3 in the tracing threads:
http://www.diystompboxes.com/smfforum/index.php?topic=48469.0
http://www.diystompboxes.com/smfforum/index.php?topic=57936.0
Is it at all possible that this is supposed to be a 500pF (470pF) instead of 50pF (47pF) cap? Would make at least a bit more sense as a radio frequency filter in this strange position.

For reference: I am still referring to this schematic:
(http://www.basicaudio.net/Boss-Tone-tech.jpg)


Thanks and Cheers,
Andy

P.s.: Sorry to those who oppose resurrecting old threads but in this case I think it's useful to have all the info in one place.

Have you figured out the Q1 Q2 connection?

Taxonomy aside, this is a VERY high-gain stage with two similar HF poles. It loves to oscillate MHz.

50p seems small to me, but if it works don't mess.

> a common collector stage

No.

QuoteBut there is something that I just now realized I don't understand: What the actual funk does C3 do? If Q2 were an NPN and upside down, then C3 would be between Base and Collector and make perfect sense as a treble cut on Q2 (as seems to be claimed or insinuated in all threads on the topic I found here and elsewhere).

The value of C3 came up about 15 years ago on this forum, someone reported it was in fact 47pF (and the other ceramics were 20n).  It does bugger-all to the tone.   

I have a feeling it might have been added to stop the thing oscillating as it helps roll the gain off to a safe level in the same frequency region that stray capacitance would cause oscillation.  Maybe not the best way to do that but it is what it is.  Looks like even 1n would work.

[IIRC, problems with that effect inspired Aron to start this group.]

Quotea common collector stage

QuoteNo.

It presents a larger collector resistor to Q1, which increases the gain.   You wouldn't be able to get such a high gain without it.  Note the DC biasing is still "good".

Hi Paul, hi Rob,

Quote
> a common collector stage

No.

No? Why not? Since the Collector is common to Q2's input and output, I thought that would make it a Common Collector, regardless of the role the transistor plays otherwise. Well, maybe that's not how the nomenclature works, then.

Quote50p seems small to me, but if it works don't mess.

Is that a dare ;) Challenge accepted...

Anyway, here is how I think the Q1-Q2 arrangement of the Bosstone works, please correct me where I'm wrong:
If we take out Q2 and R2 the whole thing is a completely normal Common Emitter stage with voltage divider bias (with AC cut for higher gain and impedance but that's not important here) and clipping diodes at the end (also not important). Almost like a Linear Power Booster with Electra Distortion clipping. Putting Q2 and R2 back in, we bootstrap this thing by moving the positive supply at the R2-R6 junction up and down in unison with the output of Q1.
When B_Q1 goes low, Q1 closes, C_Q1 goes high. Now Q2 tries to keep its Emitter and therefore the R2-R6 junction at the same voltage as C_Q1 by closing.
When B_Q1 goes high, Q1 opens, C_Q1 goes low. Now Q2 must also open to pull the R2-R6 junction down.
Is that so? Please enlighten me with the correct terminology for this sort of thing.

If my analysis is correct and if C3 is supposed to kill oscillations, wouldn't it make more sense to make C3 go from the R2-R6 junction to ground, or in the negative feedback of Q1, or from the R2-R6 junction to B_Q1, or from B_Q1 to ground? I have had some problems with oscillations on the breadboard despite a 68p mica cap as C3. Sticking in a 3n9 cap parallel with the diodes (which is the same as from the R2-R6 junction to ground for high frequencies) took care of that without noticeably changing the sound. I'm just wondering if the placement of C3 in the original is one of these "it sort of works well enough so lets not think about it anymore" things that could actually be done better. Since many people report oscillation problems with the Bosstone, I would think that this is not as moot a discussion as it might seem at first.

Now that I spell it out like that, this also explains why this circuit reacts so well to replacing R2 with a (100k-250k) pot since R2 controls how effectively Q2 can do it's thang on the up or down swing.

QuoteIt presents a larger collector resistor to Q1, which increases the gain.   You wouldn't be able to get such a high gain without it.  Note the DC biasing is still "good".

I'm afraid I don't quite follow. Could you elaborate on the "larger Collector resistor" analogy/explanation? It "feels" like this might be a different way to look at what I described above but it seems to me that the "virtual Collector resistor" made up of R2, R6 and Q2 is not merely large but changes value in accordance with the signal. Also: What is "good" DC biasing, in this case and why wouldn't it be good?

QuoteI have a feeling it might have been added to stop the thing oscillating as it helps roll the gain off to a safe level in the same frequency region that stray capacitance would cause oscillation.  Maybe not the best way to do that but it is what it is. Looks like even 1n would work.

Is it just me, or does the Bosstone have more post sentences starting "I have a feeling it might..." than most other effects? This really is a quite unique beast and I wonder why. It sounds fantastic and is enormously versatile yet it seems to have very few descendants expanding on this topology. There are clones galore but they mostly just plain copy the old original (often right down to using ceramic caps and carbon resistors) or make modest changes rather then developing upon it as has been done very extensively with Fuzz Faces and Big Muffs. Curious, that, isn't it? Lack of a "Technology of the Jordan Bosstone" article, maybe?

Cheers and thanks,
Andy

Quote from: Fancy Lime on December 12, 2017, 06:51:48 AM
No? Why not? Since the Collector is common to Q2's input and output, I thought that would make it a Common Collector, regardless of the role the transistor plays otherwise. Well, maybe that's not how the nomenclature works, then.

Quite right (and quite wrong)..

If you consider Q2 stage as "stand along" you could name it CC but if you consider it as a "load" to Q1 (roughly h_FE times R2) you can see the role of C3 as Q1 Collector -split resistor- feedback capacitor..

Quote from: Fancy Lime on December 12, 2017, 06:51:48 AM
What is "good" DC biasing, in this case and why wouldn't it be good?

Ahaaaaa...!!  :icon_biggrin:
(Any biasing better than another "bad" one should be consider "good"..)

To be more serious, in such a high gain arrangement any Collector current variation (other than nornally expected due to circuit fix set gain) should result in respective Gain non-linearity (distortion) in a non musical way..
(waveform curve shouldn't be a "smooth" line..)

QuoteTo be more serious, in such a high gain arrangement any Collector current variation (other than nornally expected due to circuit fix set gain) should result in respective Gain non-linearity (distortion) in a non musical way..
(waveform curve shouldn't be a "smooth" line..)

Well, the Bosstone is not exactly a HiFi phono preamp, is it? And isn't "gain non-linearity" sort of the point of a fuzz?

Back to the C3 mystery:
So, I swapped around some parts and fiddled with the knobs to see if I could get it to oscillate so I could try different countermeasures. Without C3 I quickly managed to get a high-pitched squeal that was tunable with the gain knob and some Russian talk radio. Putting a 68nF mica cap as C3 in it's traditional space made absolutely no difference whatsoever. Neither did replacing it with a 1nF cap. A 1nF cap from B_Q1 to ground killed the radio but not the squeal. The 68nF mica between B_Q1 and C_Q1 brought the silence: no squeal, no radio. I cannot hear any difference in sound other than the lack of noise even in clean settings (lowest gain of the modified gain control and no clipping diodes), which I find a bit surprising, considering the relatively large value and the large 560k+560k high side bias resistors. Guess I found my setting then.

Cheers,
Andy

Quote from: Fancy Lime on December 12, 2017, 10:26:56 AM
And isn't "gain non-linearity" sort of the point of a fuzz?

If you presume so, change your query about "good biasing" to a more apropriate like "fuzzy biasing"..  :icon_biggrin:

My bad about previous note:  :icon_redface:
C3 actually by-passes part of Q1 Collector resistor, redusing Gain (and oscillation) in high frequencies..
(doesn't act as "classic" feedback cap..)

Hi,

so here's my current variant:
(https://s17.postimg.org/pt7w6k7uz/Bosstone_Variant.png) (https://postimg.org/image/pt7w6k7uz/)

For the most part a standard Nashville Bosstone plus Mark Hammers Stupidly Wonderful Tone Control. The clipping diodes are switchable because this thing sounds nice, different and interesting with 0, 1 or 2 diodes. 1N4001's seem to sound a bit grittier than 1N4148's, which I attribute to them being a bit slower but that may be imagination. LED's are a bit of a middle ground between the 1N4001's and no diodes. The Gain control is now implemented as a variable Emitter resistor AC bypass for better noise characteristics in combination with the MPSA18, which worked better than I expected. I can no longer make out any self noise of this thing through my already very quiet headphone amp. Nice. The feedback-cut cap (now C3) is switchable because I was missing some better low gain sounds from the original design. It is  connected via Schottky diodes, which I originally added as a noise gate on my last build where I used the traditional pre-gain control and a 2N2222 as Q1. With the new Q1 this is no longer necessary for the self noise but still kills any hum coming from the guitar, which is nice in extreme gain settings, without affecting even the softest picked notes. One could make the Boost switch an on/off/on type and have it both ways, if one wanted but for me there are no downsides to having the diodes although the benefit is not as great anymore as it used to be. That's pretty much it. Oh yeah, proper power supply filtering seemed like a good idea because of the insane amount of gain available. And of course a metal can for Q2 for extra style points ;)

Tone-wise this is by far the most versatile fuzz I know. If I was only allowed one fuzz, this would be it, no discussion. Goes from fat raunchy "transistor overdrive" through the classic 60's tones (think, Acoustic 360 fuzz, Orpheum, Mosrite, Shin-Ei Companion), most of the better Big Muff tones and high gain metal distortion all the way to insanely compressed, implosive duck-and-swell sounds that fold in on them selves before expanding and filling the room. And every combination of those.

One question though: Why does this thing produce an octave down effect in some settings (high gain setting, medium-high impact, both diodes in)? It is actually a really cool synth-organ-like effect and tracks better than some "real" octave down effects. But why? And how? I've read about this phenomenon before (I think Mark wrote something about it) but never heard an explanation.

Cheers,
Andy

QuoteI'm afraid I don't quite follow. Could you elaborate on the "larger Collector resistor" analogy/explanation?

There's a few ways to view it, all of which are equivalent.  The easiest and most intuitive way to understand it is to think of the two 18k's and Q2 as a bootstrap connection.  This thinking also gives you an idea of how the designer come-up with the otherwise obscure connection.  If Q2 was an NPN boot-strapped common-collector stage it would have a bootstrapping cap from the output back to the tap point between the two resistors.  Using the PNP lets you remove the cap.  Amplifiers using boot-strapped collector loads to get more gain are quite common, in fact a large percentage of power amplifier use this idea in the main gain stage.

When I said "good" biasing I meant the DC at the collector of  Q1 is near mid-rail.   If you tried to build a CE amp with such a large collector resistor it would operate with a very low collector voltage and have little output swing.

QuoteA 1nF cap from BQ1 to ground killed

For oscillation problems it's sometimes a good idea to have a smallish resistor in series with the cap.  The resistor allows the high frequency gain to be reduced but has less HF phase shift, which makes it less prone to oscillation. Tweaking two parts by hand is harder but is still doable.

QuoteGuess I found my setting then.

68n is probably audible in that it will roll-off some highs.

With RF and oscillation issues caused by stray capacitance, often a small cap across the input, perhaps after the 100k pot, is best.   For this ckt a small cap across the output might help too.  Ideally one cap should dominate in the sense that it just affects the audio and the other is a "helper" for the HF zone; sometimes using both caps can cause issues.

You want to tame the HF region without affecting the audio region.

QuotePutting a 68nF mica cap as C3 in it's traditional space made absolutely no difference whatsoever.

FYI:
- 1n is the point where you might hear something.
- Any more than about 2.2n appears to achieve nothing as far as suppressing oscillations.
- Above 10nF starts to make a band-pass filter.   68n  produces a band pass with very large narrow peak; quite angry looking.

keeping an eye on this thread as im currently venturing again into boss tone territory.  its a fun one. will post when I've got something nailed down.

> Collector is common to Q2's input and output

Input appears as a floating voltage across R4, Q2 B-E. There is no NFB as in an emitter follower.

Q2 appears funny because the load is connected at the emitter. But look at *loops*. Q2, R5 etc, and power supply is a loop. It works the same in any order. Q2 is common-emitter.

Two CE stages is VERY high gain. It wants to oscillate. The 50pFd is correctly placed to shift one of the poles and reduce trouble.

QuoteQ2 is common-emitter.
Two CE stages is VERY high gain..

I "have to" (cough) disagree.

If that was the case the signal gain from the input upto the collector of Q1 (=base of Q2) would only represent part of the total gain.   In reality the collector of Q1 shows *all* of the gain.  Q2 is in fact a CC amp but it's connected as a bootstrap.

The common form is like this, (I'm sure you have seen it)
http://www.aqpl43.dsl.pipex.com/ampins/discrete/2Q-VEM/2Q-VEM%20pic1.gif

I know the mechanism/trick you are talking about.  That view works when you analyse a one-transistor Colpitts oscillator; You make the emitter the "ground" and take it from there.   I don't think you can do that here because the input and output voltages are referenced to a common ground.

Imma throw some gasoline on the fire and say Q2 is a DC connected buffer (emitter follower) using Q1s collector voltage for bias.

:icon_biggrin:

QuoteImma throw some gasoline on the fire and say Q2 is a DC connected buffer (emitter follower) using Q1s collector voltage for bias.

It's kind of that.   That view would make the effective collector resistance of Q1 (hfe+1) * RE.

However,  the resistor RB between the Base and Emitter of Q2 takes some of the gain away so you end-up with the impedance looking into the RB +Q2 as, 

      Zin = (hfe + 1)  RE RB / (RB + Rpi)

which is less than (hfe+1) * RE.

Rpi is Q2's hybrid-pi resistance.   We can simplify a bit by realizing Rpi depends on the Q2's collector bias current.  The collector current is set by the bias point voltage across RE , ie. I_RE = (9-VE)/RE.   The collector current is about 84% of I_RE because the bias current through RB (=VBE/RB) steals some of the RE current.   If you crunch the numbers you end up with Rpi  approximately 1.8 * RE.

So that means,

            Zin  = (hfe + 1) RE RB / (RB + 1.8 RE)

If we now choose RB = RE as per the circuit (the 18k's)

          Zin = (hfe + 1) RE / 2.8

So the presence of RB  reduces the gain by a factor of 1/2.8 from the straight DC connected buffer view.

Here's an AC coupled version,  (R2 on this schematic has no equivalent on the Boss Tone)
http://www.nutsvolts.com/uploads/wygwam/NV_0903_Marston_FIG19.jpg
From,
http://www.nutsvolts.com/magazine/article/bipolar_transistor_cookbook_part_3
(bit more here http://www.nutsvolts.com/magazine/article/bipolar_transistor_cookbook_part_2)

> The common form is like this

That also is CE-CE. Q2's input is just Vr4, B to E.

> the collector of Q1 shows *all* of the gain.

Because it bootstraps from Q2. But the input to Q2 is just the ~~20mV at Q2's B-E port.

QuoteThat also is CE-CE. Q2's input is just Vr4, B to E.

Sure, all transistors, independent of the circuit configuration, are transconductance devices.  So they all have an input between B and E.

transconductance  doesn't mean CE.   CE and CC are special cases.

When you use the term CE amplifier  the inputs *and* outputs of the circuit must use the "E" as the common ground reference; that's why it's called *Common* Emitter.   The concept of gain only means something with that constraint.  Where there are external input and output connections these have a pre-defined ground node.  You are not free to choose a different ground node if you want to call the connection a CE amplifier.   If you have two CE amplifiers the emitters must share the same "Common" ground (the power rail obviously being treated as ground for ac signals as well.).     Only then can you multiple the gains of each stage to get the overall gain.  If they don't share a common ground then one or none of those stages is allowed to be called a CE amplifier and the gains don't multiply.  Sure, they are still a connection of transconductance amplfiers but they are not a cascade of two CE amplifiers.

For the case of a one transistor Colpitts oscillator there are no external connections so  we are free to move the ground node to E.

I can give two examples where the problem is clear.

1) A CC amplifier ie a buffer.    The gain is 1 (well just under).  We can measure the input and output voltages referenced to ground, which is the collector.  The reason we call this CC is *because* C is the common ground connection.

Suppose we have a 9V rail and a RE=10k and the output is biased to 4.5V  then the transistor's small signal re is approximately 60 ohms.   If we take the view that this is a CE amplifier, with the ground at E, the gain ~ 10k/60 ~170 from the input between B & E and the output between C and E.   *BUT* because the external connections have their grounds referenced to C and the input to output gain is 1 not 170.

2) Now consider the case of connecting two CC amplifier in series.    Here the gain is still 1.  If we view one of these as a CE amplifier we would choose one of the E's as the ground node.   The first issue is the E is not the ground node for the input and output signals but say we ignore that.  The E of the first stage is different from the second stage.  There's no way we can draw the circuit  to look like two CE amplifier in cascade and so we don't see a gain of 170*170 = 28900.    If we chose the E of the other stage as ground we still get the same problem.   Each individual amplifier might have a gain of 170 from BE to CE but it is not correct to conclude that the overall gain from the input signal to output signal perspective is the product of these the individual CE gains.   

If there we no external connections, like in an oscillator, we are only allowed to redefine one of the E's to be ground.   We cannot choose one E then the other E as the ground as separate cases and compute the product of the gains.

3) I could give a third example of a CE amplifier follow by a buffer but the argument is the same as (2).

Quote from: Rob Strand on December 14, 2017, 03:04:09 AM
When you use the term CE amplifier  the inputs *and* outputs of the circuit must use the "E" as the common ground reference; that's why it's called *Common* Emitter.   The concept of gain only means something with that constraint.  Where there are external input and output connections these have a pre-defined ground node.  You are not free to choose a different ground node if you want to call the connection a CE amplifier.

Correct me if I'm wrong, Rob but that only stands for grounded Emitter CE amp..
(i.e. not in case of Emitter degenerated amp w/o Emitter by-pass cap..)

Unless you refer to "ground" as the point standing to the greatest extend from Vcc (something like orientation of JFET's Source pin ..) but in such a case it should be possible to exist some "different" value GNDs..

QuoteCorrect me if I'm wrong, Rob but that only stands for grounded Emitter CE amp..
(i.e. not in case of Emitter degenerated amp w/o Emitter by-pass cap..)

Yes you are correct.    I was yacking about the high gain case.  It makes the main point easier to get through.   The degenerated case is the same idea (since RE just adds onto re and reduces the transconductance) but then you have to keep saying "or the ground side of the emitter resistor".

OK, now...
(gasoline jerrycan holder seeking for matchbook owner..) :icon_redface:

Despite its designation, does Q2 "add" gain or not..??  :icon_mrgreen:

Being more of a nuts-and-bolts guy, let me sneak in here to ask if adding the 68pF B to C of Q1 is the cure for the noise issue(s)?  I put together a layout for this recently, and am glad to have run across this thread.  Larry

Quote from: rutabaga bob on December 14, 2017, 09:50:23 AM
Being more of a nuts-and-bolts guy, let me sneak in here to ask if adding the 68pF B to C of Q1 is the cure for the noise issue(s)?  I put together a layout for this recently, and am glad to have run across this thread.  Larry

Almost. If you look at the last schematic I posted on this thread you can see a 3n9 cap (C6) across the diodes. This C6 is necessary at minimum settings of the Tone control to stop some bizarre low-frequency oscillations at high Gain / high Impact settings, which may or may not be a breadboard-specific problem due to parasitic capacitances and crosstalk between neighboring tracks. But even without a tone control, a small cap across clipping diodes is often a good idea to keep the clipping musical and not overly harsh and it also helps keep high-frequency noise down. That aside, the 68p cap across B-C of Q1 kills all oscillation tendencies in my breadboard unit (which for this beast may or may not mean that it will do the same for your build but it's a starting point). 68p is a bit on the large side and actually cuts of at about 4.2kHz. I could not here a difference through the fuzz even at minimum gain settings without the clipping diodes with a bass. If you intend to feed a Tele into this, 47p (6kHz) or 33p (8.6kHz) may be more suitable, I just did not have mica caps with those values and I have not jet tried ceramics. Ceramics are probably fine, I was just going for a "lowest possible noise" version and ceramic caps are supposedly "noisy" and microphonic. Supposedly. Or allegedly, which seems to be the Word of 2017.

Cheers,
Andy

Thanks!  I was planning to do a 'regular' version to start.  Cheers!

Hi Larry,

the original version is awesome as it is but I would most certainly encourage you to breadboard this thing and try some of the extra bells and whistles. Makes this by far the most versatile fuzz pedal I know of and is just soooooo much fun to play around with. There are settings that sound like a pretty convincing tube amp emulator. Thats not the point of this pedal at all, just goes to show how insanely versatile it is for such a simple circuit.

Have fun,
Andy

QuoteDespite its designation, does Q2 "add" gain or not..??

No it's just a buffer (so gain = 0.9xxx ~ 1.0) .   The signal at the base pretty much identical to the signal at the emitter.

Putting the finer arguments aside, here's my summary of the key points,
- Q2 acts a buffer.   The signal at the Q2's base is roughly equal to that at the Q2's emitter.
- The collector load for Q1, which determines the gain of Q1, is roughly RE (the emitter resistor of Q2) times the gain of Q2; as Groovenut mentioned.  However it's actually less than that due to the presence of RB (the base-emitter resistor of Q2), roughly half;  see my response to Groovenut.
- RB sets the bias current of Q1 to a sensible value.   IC of Q1 = VBE of Q2  / RB.   If RB wasn't there the collector current of Q1 would be very small.
- The bias voltage at the output is set by the 2x560k's and the 150k.
==================================
We can take things one step further.  The re of Q1 is determined by IC of Q1,

         re1   =   VT  / (VBE / RB)     = RB * (VT/VBE) ;    where VT = 26mV and VBE ~ 0.65V

The gain of Q1 is then (CE amplifier gain)

      gain = collector load of Q1  /   re1

The collector load of Q1 is the "Zin" I gave before

       Zin = (hfe + 1) RE RB / (RB + 1.8 RE)             ; hfe = gain of Q2

So
       gain = (hfe + 1) (VBE/VT)  RE / (RB + 1.8 RE)

If RE = RB then

       gain = (hfe + 1) (VBE/VT)  ( 1/2.8 )

              =  (0.65/0.026) * ( 1/2.8 ) * (hfe + 1)

              =  8.9 * (hfe + 1)

So when RE = RB, the gain is roughly 9 times the hfe of Q2.   
Which is 60dB to 70dB.

The actual gain is a bit less because input impedance of Q1 loads the pickup.

The gain of a CE amp running at a 4.5V collector voltage is  CE_gain = (4.5/VT)  = (4.5 / 0.026) = 170 or 45dB.
If we had a straight buffer after it the gain would still be 45dB. 

So the tricky connection of the Boss Tone gives you an extra 20dB gain.

Exactly Andy, i was playing with mine, modified as per jimi photon, and with a epi sheraton and 5watt epi, i was getting some electric mud alike tones, not that i play that good, but sound was there. Impressive

> For the case of a one transistor Colpitts oscillator there are no external connections so  we are free to move the ground node to E.

It is not about "ground". Transistor does not know ground from a hole in the dirt.

It is about what is "common" to the input and output *loops*.

First-- we may simply be whipping both sides of the same dead horse, and denying the other's whippage.

Second-- that drawing is all bent up. Let's draw it out clearer.
(https://s33.postimg.org/ewyzjvv23/CECE.gif) (https://postimg.org/image/ewyzjvv23/)

Nothing is changed topologically. It may be mildly clearer that Q1's output appears only as a floating voltage source (my)R127. Q1's intrinsic collector impedance is infinite (about 2Meg), so both ends of R127 can float to any voltage. This is constrained by the top of R127 being tied to Q2 E. R127's voltage appears directly across Q2 B-E.

Q2 is also perverse in that the usual sequence of battery and resistor has the battery and resistor swapped. I think Broskie refers to "chain of pearls"(?), that you can slide these things around and the *loop* still works the same.

I marked-up where Q2's In and Out loops come together. The "common" is the Emitter.

For self-gratification I plotted the gain vs. frequency curves, for Q1 B-E, Q2 B-E, and Q2's load resistor. Oddly the 50pFd has about no effect. While the CE-CE is prone to oscillate at unity-gain, here we are far from unity gain. Also the two transistors are at very different currents, and their working fTs are about a decade apart (Q1 6MHz, Q2 60MHz). The 50p may be "a good idea" which is not needed.
(https://s33.postimg.org/61y59dlp7/Boss_Tone_FR.gif) (https://postimg.org/image/61y59dlp7/)

The gain of Q1 is about 10, which we would have expected from Shockley. The gain of Q2 is near 80, again validating Shockley. Total gain is near 750.

For comparison, I fotochopped-in the connection to make Q2 work as a follower, while keeping DC currents the same. Gain of Q1 rises to 170, Q2 now works unity gain, total 170. However you account this, the BossTone has about 4X the gain that we would expect from Q2 being unity gain.

I do not know why it bumps at ~400Hz; I assume the two poles through the DC feedback network. I threw-in a source impedance (there are no infinitely strong audio sources) and get significant fall-off above 3KHz, much of this right at Q1 Base. There is no real sign of a 10MHz bump, the usual trouble with a Sziklai (which this is, except no overall NFB).

Coming around to your side of the horse: yes, this can be studied as a bootstrap. And wow, it comes to the same answers (math is funny like that). I feel the CE-CE analysis is more direct and insightful. I know you are a very sharp thinker and if you still disagree, oh well.

As the saying goes: "The only major difference between magic and science/engineering is understanding how it works." I start to understand why the Bosstone in particular is surrounded by a misty aura of magic. I wonder how this thing was first designed. Seems hard to imagine that the designer came up with this on a piece of paper, thinking it was an obvious solution and then no-one ever again though the same way. Or did they just accidentally stick an PNP where an NPN should have gone and found it worked and went on refining from there? The California version is a whole lot more conventional, and when you take that, replace the NPN Q2 with a PNP, ditch the cap between the stages and the n-feedback resistor of Q2, you arrive at a topology very similar to the Nashville version.

But apart from historical musings: It makes sense to me that "commonality of loops" should be the decisive factor for terminology. Although the way I had it explained to me was: The terminal, which is not the input and not the output is "common" to both. Therefore, if the input is the Base and the output is taken off the Emitter, it is a Common Collector. Which also seems to make sense. I also learned that (simplified and always assuming B is the input) the Emitter output gives us current gain (at ~voltage unity) and the Collector gives us voltage gain. Making a CC a buffer and a CE a voltage gain stage after the above mentioned naming scheme. Can someone clarify? Also, if the loops are the decisive factor, what makes and what breaks a loop? Working backwards from Paul's last post it seems that rails and active elements break a loop, correct or not?

Thanks,
Andy

Quote from: Rob Strand on December 14, 2017, 04:28:54 PM
The gain of a CE amp running at a 4.5V collector voltage is  CE_gain = (4.5/VT)  = (4.5 / 0.026) = 170 or 45dB.

Or 180 (half a db higher), if we take VT=25mV (which, IMHO, is closer to reality 'cause 26mV are considered for 20^oC working temperarure..) confirming the rule-of-thumb for maximum Gain of a CE(grounded & unloaded) amp, biased at V_CC/2, being 20 times V_CC..  :icon_biggrin:

P.S.
Part of my previous post had to do with putting on fire with gasoline (which, I think, worked well..  :icon_redface: ) and part of it for checking if Q2 arrangement is set to play some "active load" role..

But, different circuit analysis points of view don't have to be identical to be both correct..  :icon_wink:

Andy...
  Doesn't that .0039uF cap across the output cut an awful lot of high end?  That's 3900pF...the stuff I've seen before is in the 150 - 220pF range.  Just asking...

Hi Larry,

I wouldn't say it cuts an awful lot of high end, just a bit of awful high end ;) Kidding aside, a cap in this position is not really part of a "proper" R-C circuit and therefore does not form a normal low pass unless you put a resistor before it because the only resistance before it is the output impedance of Q2, which seems to be rather small (but at this point in the thread I hardly know what's what anymore when it comes to them dang transistors). That is why the Stupidly Wonderful Tone Control works really well in this circuit. When the 47n cap C7 is shifted all the way to the input of the Tone pot, it cuts almost no treble, shifted to the output, it forms an R-C circuit with the (at first parts of and then the full) 5k resistance of the pot. The 3n9 cap is mostly there for killing oscillations, I cannot really hear a difference when I take it out or put it back in (in the settings where it does not oscillate anyway). That being said, try smaller values if you like, it will probably work. 3n9 was simply the first thing I happened to stick in there and it worked exactly as I hoped so I did not fiddle with it. But I'm pretty sure this is not a critical value.

Cheers,
Andy

QuoteFirst-- we may simply be whipping both sides of the same dead horse, and denying the other's whippage.

Thanks for putting the response together.
What you have done there is a little different  to what I thought you meant before.
I understand what you have done here 100%.  I've no disagreement.
In fact I've used that idea many times before.

For your analysis (case A) the output of the first stage, and the input of the second stage is
defined as vb2-ve2.

         gain1A    =  (vb2 - ve2) / vin    = 10    ; first stage
         gain2A    =   vout / (vb2 - ve2) =  80   ; second stage
         gainA        = gain1A * gain2A    =  vout/vin

When we calculate the overall gain the internal voltage (vb2 - ve2) cancels out.

For my analysis (case B)  the output of the first stage, and the input of the second stage is
defined as vb2 (referenced to ground).

         gain1B    =  vb2 / vin     = 800     ; first stage    (I actually got  about 900 but 800 is OK for the argument)
         gain2B    =   vout / vb2  = 1.0     ; second stage
         gainB        = gain1B * gain2B    =  vout/vin

When we calculate the overall gain the internal voltage vb2 cancels out.

Obviously the overall gain agrees gainA  = gainB.

However if we want to ask the questions: What is the gain of the first stage? and what is the gain of the second stage?  We end-up with a dilemma  because your first stage gain is small and my first stage gain is large.
The dilemma obviously arises because of the different definitions of what the output voltage of the first stage is.

Suppose we try to answer these questions by measurement.  We take a CRO and measure the input voltage, the collector voltage of Q1 and the output voltage, then calculate the gains of the first and second stages.

The numbers would match my case B values.   That doesn't mean the case A values are wrong.  It's just that the definition of the case A  gains don't match how we *normally* consider gain to be defined.   

If I chose to measure the voltage  (vb2 - ve2) and compute the gains from that then we would end-up with the case A gains.   The thing is we don't normally consider the gains measured like that.   (There are cases where you *have* to measure gains like that as it has no other meaning.)

We can ask the circuit what it thinks the gain of Q1 is by adding a capacitor, say Cf = 33pF, between the collector and base of Q1.    We then look at the input capacitance of Q1 and see what it is, alternative we could add 10k series resistance and measure the -3dB point and deduce the input capacitance.   The Miller effect will multiply this capacitance by the gain of the first stage (Q1),  so

      Cin   = (gain1-1)  * Cf

From Cin and Cf we can find what the circuit sees as gain1.

If we do this what we find is Cin corresponds to the large gain of gain1B  not the smaller gain of gain1A.

The analysis of case A can be used to derive gain1B.   We simply divide the total gain gainA by 1, the gain of the output buffer, and use that value for gain1.   Which is in fact identical to gain1B!

QuoteComing around to your side of the horse: yes, this can be studied as a bootstrap. And wow, it comes to the same answers (math is funny like that). I feel the CE-CE analysis is more direct and insightful. I know you are a very sharp thinker and if you still disagree, oh well.

There's nothing wrong with your analysis.  I have no disagreement.    I even agree that the method is often easier for calculations.  I use different methods depending of what I want to know.   The *only* disagreement, and it's not even a disagreement, is the definitions of gains in each case is different and in your case it's not what we normally use when we measure the gain of a stage.  For analysis we don't care what internal variables are, they are just an ends to means.

QuoteOr 180 (half a db higher), if we take VT=25mV (which, IMHO, is closer to reality 'cause 26mV are considered for 20oC working temperarure..) confirming the rule-of-thumb for maximum Gain of a CE(grounded & unloaded) amp, biased at VCC/2, being 20 times VCC..  :icon_biggrin:

For an office I think it's 25.5mV, I often use 26mV because it's between that and the 300K value PSPICE defaults to.

There's more fine grain problems with that circuit the 'ro' value of the transistor Q1 has an effect on the gain, maybe 10%.  If you look at Ic1/vbe1  it doesn't agree with the gm value based on the Q1's collector current as some output current is lost through ro.

QuoteBut, different circuit analysis points of view don't have to be identical to be both correct..

Indeed!

QuoteSeems hard to imagine that the designer came up with this on a piece of paper, thinking it was an obvious solution and then no-one ever again though the same way.

How you come-up with these things depends on your mind set.

I see it as a bootstrap technique - very common way of getting more gain.
Groovenut's post is an equally valid way to start the idea, as a second step you then add RB to fix the bias current.

> I wonder how this thing was first designed.

Early transistor writings listed zoo-fuls of possible connections. From Cowles (https://www.amazon.com/Analysis-Design-Transistor-Circuits-Lawrence/dp/0442017103):

(https://s33.postimg.org/5utumfc5r/Cowles-_Fig10.1.gif)

Here is how I see the AC equivalent. "Practical" details like batteries, bias, ground omitted.

(https://s33.postimg.org/ffdh9bwcv/ded-horse.gif)

QuoteEarly transistor writings listed zoo-fuls of possible connections. From Cowles:

Interestingly if you take fig 10.3 and change the second stage from a PNP CE  stage to an NPN CE stage you get about 20dB more gain (because it forces Q1 to operate at a lower collector voltage).   That pattern is essentially the Fuzz Face pattern (although Fuzz Face has a lot more biasing baggage.)

> if you take fig 10.3 ...... essentially the Fuzz Face ....has a lot more biasing baggage.)

That collage omitted biasing.

Consider fig 10.4:
(https://s33.postimg.org/lihq79p3f/Cowles-_Fig10.4.gif) (https://postimg.org/image/lihq79p3f/)

QuoteThat collage omitted biasing.

Consider fig 10.4:

Thanks, very cool.

I remembered I posted this some time ago,
http://www.diystompboxes.com/smfforum/index.php?topic=110917.0

The Bosstone circuit presented at the start of this thread implements a typical shunt-shunt global feedback configuration over Q1 and Q2, where the feedback network (560k + 0.022uF + 560k) moves from closed to open loop along with growing frequency through C2. The schematic below shows my simplified analysis version, where the simulation model is on the left side and the small-signal model on the right side:

(http://www.guitarscience.net/img/shuntshunt2.png)

From the basic feedback theory, the gain for the shown circuit in closed loop (low frequency limit) is (RB1a+RB1b)/RS = (560k+560k)/10k => 40 dB. In open loop (high frequency limit) the total gain is determined by Q1 as -gm1*RL, where gm1 is the transconductance of Q1 and RL is the effective load on the collector of Q1. Since the input impedance of the "emitter follower buffer Q2" is huge, the RL in open loop is approximately 560k (RB1b), since the feedback loop is split into input and output sides via grounded C2 and RB1b appears parallel to the huge buffer impedance. Here is my simulation on the simplified circuit, showing the frequency response of Vout/Vin:

(http://www.guitarscience.net/img/freqresp.png)

Note that the second 18k resistor affects mainly on the biasing of Q1, it does not have much significance on the gain as it is parallel to rpi2 in the small signal model. Similarly the effect of the "magical 50pF" cap is lost, because the BJT internal capacitances are most likely higher that this, resulting in a natural HF roll-off.

If someone is experiencing oscillations in this circuit, I would personally try to add a small (maybe 220 ohm) emitter resistor to Q1. This will add local negative feedback, and hopefully stabilize the high gain response of Q1.

I see the Q2 only as a buffer in this circuit. For the loop gain in closed loop state it contributes as a significant gain multiplier hfe2+1, and in the open loop it presents a high-impedance load to Q1.

I've used a 100 ohm on q1 emitter to calm these down, seems to work. Just copied the idea from a Fuzz Face modification. Lower gain transistor in q1helped too, and a series resistor on the input as well. Tried a lot of things, funny beasts, some are fine right out of the blocks, others can be a pain to sort. Great fuzz though.

> I see the Q2 only as a buffer in this circuit.

Buffering what? Unity gain to what?

The collector impedance of Q1 is "infinite". I believe it is >10X any other impedance in the area. So the collector of Q1 is not some kind of "voltage source" that we can "unity gain" buffer.

This is however semantics. Except in that one interpretation or the other can, with reasonable simplifications, lead to incorrect answers.

I seem to have lost my sim but your gain results look similar to mine. I agree the 50pFd may be pointless, the kind of thing audio designers throw-in "just in case".

Quote from: PRR on December 24, 2017, 11:48:56 PM
The collector impedance of Q1 is "infinite". I believe it is >10X any other impedance in the area. So the collector of Q1 is not some kind of "voltage source" that we can "unity gain" buffer.

This is correct, the collector impedance of Q1 is not only any single resistor value as I first thought it would be in the open loop state.

Quote from: PRR on December 24, 2017, 11:48:56 PM
Buffering what? Unity gain to what?

When referring to Q2 as a buffer, I was thinking more from impedance point of view than gain. For some of us it is easier to see it that way.

> the collector impedance of Q1 is not only any single resistor value

A "clean process" (post-1972) BJT has collector impedance about 10,000X the emitter impedance, At 30uA, pencil near 1K at emitter and near 10Meg at collector. This is very approximate; but clearly "infinite enough" compared to all other circuit values.

QuoteA "clean process" (post-1972) BJT has collector impedance about 10,000X the emitter impedance, At 30uA, pencil near 1K at emitter and near 10Meg at collector. This is very approximate; but clearly "infinite enough" compared to all other circuit values.

I think he is referring to the effective collector of Q1 as opposed to the collector impedance of Q1, 'ro'.
In my analysis I see that collector load directly.   In your analysis you get a different result because your BE impedance goes to the output not to ground.

Interestingly ro does have a noticeable effect.   The easiest way to see it is an AC analysis in PSPICE, plot IC and then plot gm1 * vbe1.  You will see IC is noticeably lower.  The missing current is explained by Q1's ro.

The early voltage is probably around the 70V to 100V zone,
https://en.wikipedia.org/wiki/Early_effect

In my analysis this comes about because Q1's ro is a significant fraction of the Q1's effective collector load.  In your analysis Q1's 'ro' appears across B and C of Q2.  Since your Q2 has a high gain ( ie. 80) the 'ro' gets significantly lowered due to the Miller effect  (it becomes 30k or so which is in the same order of magnitude as the Q2's BE impedances; rpi2~37k and RBE=18k).   As before these two views are identical.

It *is* noticeable effect although not dominant.

[Edit:  Occasionally you get data on 'ro'

See page 3 on,
http://www.datasheet.hk/view_online.php?id=1544966&file=0274\bc546a_421585.pdf

hoe = 1/ro  = Ic / VA

Take the "B" device

hoe = 30e-6 at 2mA  so the early voltage is VA  = 67V
]