Hi Guys
I'm having a go at building an smps to get decent plate voltage for a tube based pedal.
Ive tried a pre-made nixie circuit from ebay and also built a vero board design kindly posted hrere by Marcos-Munky.
https://postimg.cc/GBMv10YN
https://postimg.cc/NyGRVGym
Both circuits give good voltage but are too noisey to use. I've use a simple audio probe (coupling cap into battery amp) and a simple valve pre amp. So a couple of questions
What sort of resistance do you think would simulate a safe load?
Would I be able to filter out the noise with another inductor/resistor/cap circuit?
Has anyone had any luck with pre-made smps circuits for this application.
Thanks in advance for any help
Welcome to the forum!
I've used both of them with good results. What preamp do you want to run with one of these power supplies? You can add additional filtering if you need to.
You have tried one or both with a tube amp, and it made noise? What kind of noise? Both should be running at a frequency above human hearing.
Thanks.
I have made a version of this (which is from vigilante397's trace of a Kingsley Harlot). Only I've omited the first gain stage and gone straight from my input to plate 1 via a 68k and 1M to ground.
My idea was to drive the pedal a seperate clean boost.
(https://i.postimg.cc/DmbBWSvd/Xdtepnc.png) (https://postimg.cc/DmbBWSvd)
I got a high pitched whine (which my dog disapproved of) when using the valve circuit. I got a lower frequency buzz, maybe more like 200hz, when I tried the audio probe straight to the output of the smps.
My experience was repairing a Blackstar HT pedal. The pcb where the SMPS section was was quite charred in places and couldn't be repaired directly ( I tried and failed!). I tried using an eBay SMPS, an NCH6100HV to power the valve. This did work but, like you, I found it was noisy. In the end I built a separate pcb using the original Blackstar HT SMPS design and fitted that and it now works fine - no noise. Try looking at that if you can't get your existing SMPS to work the way you want.
(https://i.postimg.cc/qNtqs6RQ/Blackstar-HT-power-supply.png) (https://postimg.cc/qNtqs6RQ)
Or add filtering before either circuit. 100R series, 470u to 1000u to ground, possibly would help. Even do it in more than 1 section...
Quote from: Yazoo on July 11, 2023, 04:10:21 PM
My experience was repairing a Blackstar HT pedal. The pcb where the SMPS section was was quite charred in places and couldn't be repaired directly ( I tried and failed!). I tried using an eBay SMPS, an NCH6100HV to power the valve. This did work but, like you, I found it was noisy. In the end I built a separate pcb using the original Blackstar HT SMPS design and fitted that and it now works fine - no noise. Try looking at that if you can't get your existing SMPS to work the way you want.
(https://i.postimg.cc/qNtqs6RQ/Blackstar-HT-power-supply.png) (https://postimg.cc/qNtqs6RQ)
As far as I'm aware, the input supply on the Blackstar converter should be 22V *DC* (unregulated). The input diode has been added to the original circuit for reverse supply protection, not for rectification. On the Blackstar amps the input supply is AC but it is converted to DC with a rectifier and filter cap before going to the converter.
There's many discussions on this forum about the NE555 converter, as use by Marcos-Munky.
What sort of resistance do you think would simulate a safe load?
100 k ohms is typical (12AX7)
Would I be able to filter out the noise with another inductor/resistor/cap circuit?
If you have 2 or 3 stages already, the problem probably isn't in the electronics per se. Layout is quite important, especially with respect to inductors and the high current lines. Sub-harmonics can lead to them buzzing like bees, either physically or electronically.
If all else fails, the step-up converters on AliExpress are only $5.
Nixie supplies are indeed too noise because they are unregulated.
I saw a nice SMPS HV supply in Kingsley tube pedals and reverse engineered it. Made SMD and THT versions of it.
Not sure I can't get into trouble if I share the schematics but it's basicly boost topology of MC34063 using 220uH coil and IRF730A or IRF840 as a switch.
Super compact pcb. Will check later/tommorow if I can post a pic
QuoteNixie supplies are indeed too noise because they are unregulated.
The NE555 design is regulated to some degree. The transistor acts as the comparator and voltage reference.
Something not addressed yet is the use of:
- shielded inductors
- caps across the input supply
If you don't use shielded inductors there's a good chance noise will get into the audio. To some degree you can distance the inductor from the audio wiring/circuit but there's practical limits. It's possible to do a test to see if the inductor is the cause separating the converter from the audio. That assumes everything else is good. You need an input supply cap close to the converter. Also if the inductor current rating is too low the inductor can saturate and that can promote noise from the inductor.
If you don't have the input cap close the converter, then distancing the converter with long wires can result in other forms of noise from the input wiring. You need to get everything right. Keep all the layout tight around the converter: input caps, output caps, including diodes, inductor, MOSFETs - pretty much the whole thing needs to be tight. Also don't let any audio run through the noisy converter ground.
There's actually many ways converter noise to get in.
In one of the older threads I went through some details on the switch frequency and switching times. I seem to remember Marcos-Munky was trying to get more current out of the converter and that meant more attention to the details.
Well thanks very much for the feedback guys. Plenty to think about and try out.
I the meantime I've ordered one of these
(https://i.postimg.cc/9R4jxLxS/s-l1600.jpg) (https://postimg.cc/9R4jxLxS)
Quote from: snailspacejase on July 13, 2023, 05:01:34 AM
Well thanks very much for the feedback guys. Plenty to think about and try out.
I the meantime I've ordered one of these
(https://i.postimg.cc/9R4jxLxS/s-l1600.jpg) (https://postimg.cc/9R4jxLxS)
Let us know how that turns out.
Quote from: snailspacejase on July 13, 2023, 05:01:34 AM
Well thanks very much for the feedback guys. Plenty to think about and try out.
I the meantime I've ordered one of these
(https://i.postimg.cc/9R4jxLxS/s-l1600.jpg) (https://postimg.cc/9R4jxLxS)
It would be a good idea to keep that circuit away from the audio. The inductor is an E-core
which aren't magnetically shielded by nature - in fact they can have quite a bit of leakage field
which can get into the audio.
The Kingsley pedal uses a shielded inductor,
(https://i.postimg.cc/wt9HHS93/kingsley-pedal-shielded-inductor.png) (https://postimg.cc/wt9HHS93)
(https://i.postimg.cc/8J5SfMJS/radial-inductor-shielded.jpg) (https://postimg.cc/8J5SfMJS)
Ok here it goes - this has been tested and runs 1 .. 2 preamp tubes nicely.
(https://i.postimg.cc/VkRHXZW6/smps.png)
U1 MC34063AD
C1 390pF 0805
C2 2,2uF
C3 450V 8 or 10mm
R1 0805 feedback resistors
R2 need to be ~2,67k
R3 680k 0805 (depends on output voltage)
R4 10k 2W
R5 330R 0805
R6 1A fuse
L2 220uH 12x12mm
IC1 6V regulator 1A TO252
Q1 IRF730A analog SMD (IRF840)
D1 ES1J DO214AC
Here are two pcbs (currently don't have a fully populated one on hand) with a pen next to them (to give the perspective of size)
PCBs have additional 6V linear regulator for filaments that schematic does not show.
(https://i.postimg.cc/1XrYMW1J/pcb.png)
Quote from: snailspacejase on July 13, 2023, 05:01:34 AM
Well thanks very much for the feedback guys. Plenty to think about and try out.
I the meantime I've ordered one of these
(https://i.postimg.cc/9R4jxLxS/s-l1600.jpg) (https://postimg.cc/9R4jxLxS)
Should work fine but man it is huuuuuuuuuuuuuuuge.
Would be ok for a standalone preamp. For a pedal - too bulky
I find the MC34063 is also available in in 8-pin DIP from Mouser etc, so this doesn't have to be an SMD build. I'm sure the inductor could be had from there or others, too.
I've had good luck a few times w/the old 555 Nixie, but it IS sort of...dated....
The first version I built was through hole. Will take a picture if I remember some time later. Went to SMT to get smaller size. SMT devices used in this "project" are huge and can be soldered easily.
Quote from: crane on July 14, 2023, 03:59:20 AM
Ok here it goes - this has been tested and runs 1 .. 2 preamp tubes nicely.
(https://i.postimg.cc/VkRHXZW6/smps.png)
U1 MC34063AD
C1 390pF 0805
C2 2,2uF
C3 450V 8 or 10mm
R1 0805 feedback resistors
R2 need to be ~2,67k
R3 680k 0805 (depends on output voltage)
R4 10k 2W
R5 330R 0805
R6 1A fuse
L2 220uH 12x12mm
IC1 6V regulator 1A TO252
Q1 IRF730A analog SMD (IRF840)
D1 ES1J DO214AC
Here are two pcbs (currently don't have a fully populated one on hand) with a pen next to them (to give the perspective of size)
PCBs have additional 6V linear regulator for filaments that schematic does not show.
(https://i.postimg.cc/1XrYMW1J/pcb.png)
Whats c3 value?
Quote from: theehman on July 14, 2023, 07:48:07 AM
Whats c3 value?
C2 and C3 are the same - both 2.2uF
Quote from: crane on July 14, 2023, 04:10:10 AM
Quote from: snailspacejase on July 13, 2023, 05:01:34 AM
Well thanks very much for the feedback guys. Plenty to think about and try out.
I the meantime I've ordered one of these
(https://i.postimg.cc/9R4jxLxS/s-l1600.jpg) (https://postimg.cc/9R4jxLxS)
Should work fine but man it is huuuuuuuuuuuuuuuge.
Would be ok for a standalone preamp. For a pedal - too bulky
I wanted to use it in a larger pedal with three preamp tubes, I cut the fins off the heatsink to save some room. I have yet to start it as, darn it, it still takes up a lot of room.
LOL, I hope it didn't need those fins to dissipate heat well enough ;)
Quote from: printer2 on July 14, 2023, 03:18:59 PM
I cut the fins off the heatsink to save some room.
https://www.heatsinkcalculator.com/heat-sink-size-calculator.html (https://www.heatsinkcalculator.com/heat-sink-size-calculator.html)
https://celsiainc.com/resources/calculators/heat-sink-size-calculator/ (https://celsiainc.com/resources/calculators/heat-sink-size-calculator/)
https://www.allaboutcircuits.com/tools/heat-sink-calculator/ (https://www.allaboutcircuits.com/tools/heat-sink-calculator/)
]https://myheatsinks.com/calculate/plate-fin-heat-sink-calculator/[url]
(https://myheatsinks.com/calculate/plate-fin-heat-sink-calculator/%5Burl)https://coolingsourcethermal.com/heatink-size-calculator/[/url]
Quote from: GibsonGM on July 14, 2023, 04:05:13 PM
LOL, I hope it didn't need those fins to dissipate heat well enough ;)
Possibly a sign the converter is a much higher power unit.
Higher power could mean higher current pulses on the input supply => more noise!
See below.
QuoteOk here it goes - this has been tested and runs 1 .. 2 preamp tubes nicely.
Thanks for posting.
Interesting the timing cap value is 390pF. The trace I had for the through-hole version quoted 100pF to 270pF
which seemed a bit low. The 390pF (say 330pF to 470pF) seems more likely.
The SMD board has an SMD inductor. From what I can work out it's 12mm x 12mm x 8mm high. Wurth sometimes have different inductor heights for the same footprint and the inductors have different specs.
From what I can work out it's a,
Wurth (WE), PD SMT Shielded Power Inductor, 744770222
Isat 1.49A
RDC 247m ohm typ., 390m ohm max
Here's the through-hole trace from the web with a few fixes,
(https://i.postimg.cc/CRjqGhrh/kingsley-HV-converter-through-hole-sch-V11.png) (https://postimg.cc/CRjqGhrh)
There's no heatsink on the MOSFET.
Specs for OP's posted pic, aliexpress/ebay unit with heatsink.
DC-DC Boost Converter 8~32V to 45~390V High Voltage ZVS Step up Booster Module
Description:
High Voltage Boost Module
Module Properties: Non-isolated step-up module
Input Voltage: 8~32V input(the default is 10~32V input.)
Input Current: 5A (Max)
Quiescent current: 15mA (12V liter 50V, the output voltage, the higher the current will increase too quiet)
Output Voltage: +45~390V continuously adjustable (default output ±50V)
Output Current: 0.2A Max(with input, output pressure related,the higher the output voltage, output current is smaller)
Output Power: 40W (Peak 70W)
Working Temperature: -40 ~ + 85 degrees (ambient temperature is too high, please enhance heat dissipation)
Operating frequency: 75 KHz
Conversion efficiency: up to 88% (efficiency and input and output voltage, current, pressure-related)
Short circuit protection: Yes.
Over current protection: Yes. (Input current exceeds 4.5A, reducing the output voltage)
Over voltage protection: Yes. (Output voltage exceeds 410V, lowering the output voltage)
Input reverse polarity protection: Yes (non-self-healing, reverse burning fuse, try not reversed.)
Installation: Four 3 mm screws
Wiring: free welding output terminals
Size(L*W*H):60 x 50 x 20 mm
Comments:
- 40W much higher than load from tube circuits. Will the module work with the light load of the tube ckt?
- "ZVS Step up Booster Module" - perhaps this will keep noise down. Better than common MC34063/NE555
Good stuff guys.
So if I were to do a vero layout of that Kingsley through hole circuit, is the priority to keep the traces around Q1, L1 and D2 as short as possible?
I'll draw it up and get you guys to critique it if you would be so kind?
Gosh, this is all thoroughly wonderful!
I've been away from here (DIY stomp) for at least 15 years, did other stuff (art and then a few years of meditation!), but it's come back to haunt me, still don't know bugger all but I LOVE constructing!
But this thread is totally up my alley as tubes have always enthralled me and recently got a whole load of Chinese 6j1's, small signal pentodes, 7 pins as well as a good load of the Russian 6n2p eb (military 12ax7 pretty much).
Big part of that was coming across a guy in Poland, alex.hotkits@gmail.com, through facebook pedal groups, whom has pedal pcb's aplenty and, especially, gets paid through paypal then the boards (at least five) come direct from JLCPCB... I absolutely love that post modern car and share ideal!
I've a image share somewhere so I'll try and get that up and goin' as, with what's here as the boost converters theres one particular Seymour Duncan thing I've got to get goin'. SD sfx-04 twin tube mayhem (just had a quick look but couldn't find the schematic to link to but I do have a copy) and it's neato as it has both the HV requirements, for the pentodes as well us HV transistors (MPSA42 and MPSA92 set up at the input alike a fuzz face) and then a whole swathe of gyrated opamps on 15+/15- like the HM2 stuff. The sfx has the NOS tiny wee 6205 pentodes whereas I wanna use the 6J1's and the MPSA92 (PNP) cannot be found, cheaply at any rate, whereas the MPSA42 can, be found, and tipped upside down!
Quote from: Rob Strand on July 14, 2023, 08:53:06 PM
Thanks for posting.
Interesting the timing cap value is 390pF. The trace I had for the through-hole version quoted 100pF to 270pF
which seemed a bit low. The 390pF (say 330pF to 470pF) seems more likely.
The SMD board has an SMD inductor. From what I can work out it's 12mm x 12mm x 8mm high. Wurth sometimes have different inductor heights for the same footprint and the inductors have different specs.
From what I can work out it's a,
Wurth (WE), PD SMT Shielded Power Inductor, 744770222
Isat 1.49A
RDC 247m ohm typ., 390m ohm max
Here's the through-hole trace from the web with a few fixes,
(https://i.postimg.cc/CRjqGhrh/kingsley-HV-converter-through-hole-sch-V11.png) (https://postimg.cc/CRjqGhrh)
Yes - I used Wurth inductor for my build - but some other brand inductors where availible in my local "radioshack" store as well.
BTW - your schematic shows a wrong fuse value.
QuoteBTW - your schematic shows a wrong fuse value
That's what was on the web schematic. I suspected it was misread as 500mA, maybe a Fast type, but I have no info to confirm. It could be 1A.
The flyback boost converter that you have here is good for a general purpose high voltage supply but there are a lot of different topologies with different noise characteristics.
For low current, a charge pump is often seen as the way to get decent performance where you only need a few milliamps and the voltage change is not that much. There is a lot of noise since turn on and turn off of the switching device happens at maximum current.
Flyback is more efficient and switching on is done at zero current but the sudden switching off is at maximum current and this generates noise.
There is the forward converter that reduces the amount of iron needed since the output current is delivered at the same time as the input current.
There are push-pull converters of various types that can be better for noise.
There is even the current-fed converter that has poor regulation performance but it has one characteristic that is valuable in one application: if all transistors are turned on briefly due to a nuclear event, this converter keeps working because nuclear photocurrents are limited by the inductive feed and the noise is low due to the triangular current waveform which is quieter than any square wave current input.
There are resonant converters that act like sine wave transmitters where the noise output is concentrated at the resonant frequency and it is easy to design filtering for a single frequency, but some of these are also variable repetition rate where a sine wave of a single frequency is produced with some variable dead time between each sine-wave pulse.
Some converters have a fixed frequency and variable pulse width whereas others have a fixed pulse width and variable on time. It is easier to design filtering for a fixed frequency converter. You should keep in mind that simple converter designs often need complicated filtering to make the overall noise level acceptable. The flyback converter proposed here is probably good but will need an input filter that would probably be larger than the converter module and the converter should be designed for a frequency well above the audio range. If you need more than one converter in a pedal chain, the beat frequency between the converters should be either above the audio range or the converters should be synchronized.
QuoteSome converters have a fixed frequency and variable pulse width whereas others have a fixed pulse width and variable on time. It is easier to design filtering for a fixed frequency converter.
FWIW, NE555 = constant off-time variable on-time, MC34063 = constant on-time variable off-time.
For the common HV circuits kicking around there's no current limit or current feedback.
Most simple converters run in discontinuous mode so there's two off-times: an off time for when the inductor current runs dry and an off-time for the switch.
If I was going to build a tube pedal for myself, I might just use a transformer from the 120 VAC supply with a diode bridge and forget about trying to use an inverter for high voltage. There is still a possibility of noise and if you are building it commercially to sell, you need CSA or UL and in some markets, VDE approval, so the inverter is a better idea in this case. If the inverter frequency is high enough, filtering should not be that much of a problem although electrolytic capacitors lose effectiveness at frequencies above audio, so we are talking multilayer ceramic for the most predictable filtering. High frequencies require fast switching times if you want good efficiency and the usual module you can buy off the internet may not be suitable for the task. Try it and see, but you will need a filter and you must design the filter for low output impedance which means a lot of capacitance and not much inductance.
Comparing the traced Kingsley SMPS schematics from crane and Rob Strand, there is a difference in the bottom voltage divider resistor coming off pin 5 off the MC34063. crane's schematic uses 2k67, while Rob's uses 3k3. Going by the datasheet's Vout=1.25x(1+R2/R1) formula, this put's crane's version at 320V and Rob's at 259V before being dropped by the 10k resistor. This seems like a big difference to me, I wonder which one is the "correct" part to get the ~250V at the output?
Another difference is the inclusion of a diode to ground at the input. Next to the fuse it looks like an overvoltage protection diode, and if so it should be a zener with a breakdown voltage of about 15-20V? But the Kingsley part looks a lot bigger than these zeners, so I wonder what the real purpose of this diode is.
Maybe somebody has a clue about these questions? Thanks!
(https://i.postimg.cc/yJV9bD0X/smps.jpg) (https://postimg.cc/yJV9bD0X)
Quote from: Lorok on June 01, 2024, 02:56:15 AMMaybe somebody has a clue about these questions?
From what I can see crane doesn't give an output voltage. crane indicates tuning the 680k to set the output voltage. In fact you can tune either. 2.67k + 680k is going to produce a higher output voltage than 3.3k + 680k at C1. I think you get that already. (From what I can see 2.67k + 680k is going to put out 320V at C1.)
The extra variable is the 10k on the output. The voltage at C2 depends on how much current is being pulled from the load. Maybe it turns out 3k3 is a little under 250V and 2k7 is a little over 250V. The trace early on in the thread is only showing 200V for the supply. If we have an uncertainty of 200V to 250V then it's hard to know what is correct. (If we pull 2mA from the supply then that's 20V drop across the 10k, which favours the 3.3k.)
The diode is just reverse polarity protection. The diode needs to be a reasonable size and the fuse will blow when the supply is reversed.
Just some information on the 40-390V modules. Used one this week to power a 6V6 and 12AX7. The weak spot on these are suppose to be the HV diode. I blew one after drawing 300V/55mA. I replaced the diode with a pair of UF4007's and 350V, 40mA was fine. Went for broke, 390V/49mA, in free air it was about as hot as I would like. The diode was rated for 3A, the two UF4007's 2A, wondering if they used substandard parts?
Quote from: printer2 on June 01, 2024, 10:55:11 AMJust some information on the 40-390V modules. Used one this week to power a 6V6 and 12AX7. The weak spot on these are suppose to be the HV diode. I blew one after drawing 300V/55mA. I replaced the diode with a pair of UF4007's and 350V, 40mA was fine. Went for broke, 390V/49mA, in free air it was about as hot as I would like. The diode was rated for 3A, the two UF4007's 2A, wondering if they used substandard parts?
Do you know what the original part was?
Layout can promote failures as small parasitic inductances can cause HV glitches which exceed the part ratings on a small time scale and eventually cause failures. These days dodgy parts is also a possibility.
Quote from: Rob Strand on June 01, 2024, 06:25:26 PMDo you know what the original part was?
Layout can promote failures as small parasitic inductances can cause HV glitches which exceed the part ratings on a small time scale and eventually cause failures. These days dodgy parts is also a possibility.
A US3MC. I was thinking of the UF4007's getting hot and unbalance with all the other things that could go wrong. I was really surprised when it worked. Reading about counterfeit chips I would not be surprised if the US3MC either did not have the current capacity or is not as quick as it is suppose to be. The one I repaired seems stable, another one I blew and replaced the diode was bouncing around but I did not spend more time on it yet. Here is the replacements in.
(https://i.imgur.com/SQtV8Vl.jpg)
Here is a stock image. There is an additional diode in the upper right corner to give a +/- supply. I thought of taking the upper diode and paralleling it with the existing one but will it cause issues as you said? Since I have a number of the UF's I tried it and it seems ok.
(https://www.elecbee.com/image/catalog/Power-Supply-Module/ZVS-DC-DC-45V-To-390V-5A-Adjustable-Regulator-Voltage-Booster-Module-Capacitor-Charging-1307809-descriptionImage2.jpeg)
I did try something else before. I CA'd a piece of aluminum onto the body of the diode. It did give me a few more watts until it blew.
(https://i.imgur.com/F9DzQbg.jpg)
Quote from: printer2 on June 01, 2024, 07:47:22 PMA US3MC. I was thinking of the UF4007's getting hot and unbalance with all the other things that could go wrong. I was really surprised when it worked. Reading about counterfeit chips I would not be surprised if the US3MC either did not have the current capacity or is not as quick as it is suppose to be. The one I repaired seems stable, another one I blew and replaced the diode was bouncing around but I did not spend more time on it yet. Here is the replacements in.
It's not obvious to me what the cause is. I can only speculate. If you build a large number of boards and they all fail then it could be the layout or the diode batch or a design issue. If a different batch or supplier of diodes fixes it the it's the diode, otherwise perhaps it's the layout or design.
When you have one-off failure it could be anything. If you replace the diode with different diodes it might look like it's the diode type but it could be the diode and replacing it with the same diode type would also work.
Early on you mentioned it was getting hot. I'm assuming it's diode that's getting hot. That would be a significant piece of information. When you convert low voltage to high voltage with a boost converter the duty is very low and requires a high current pulse. It's possible the inductance value is too small. It could even be getting near saturation are bumping the current up.
Sorry for late reply - I don't visit this place as often as I used to. But still - here is an explanation/answer to your questions/concerns.
Input diode? Doesn't harm anything - fullfils reverse polarity protection. I think I had it on my actual build although it's missing in the schematic I have. Schematic will work without it.
Feedback voltage dividers? Different designs use different voltages. I had several of these pedals to open and they had different values. You are supposed to calculate your own values for the voltage you need. That is the beauty of it. Tested ~150....400V (unfrotunately the tests were without load) - worked OK.
Quote from: Lorok on June 01, 2024, 02:56:15 AMComparing the traced Kingsley SMPS schematics from crane and Rob Strand, there is a difference in the bottom voltage divider resistor coming off pin 5 off the MC34063. crane's schematic uses 2k67, while Rob's uses 3k3. Going by the datasheet's Vout=1.25x(1+R2/R1) formula, this put's crane's version at 320V and Rob's at 259V before being dropped by the 10k resistor. This seems like a big difference to me, I wonder which one is the "correct" part to get the ~250V at the output?
Another difference is the inclusion of a diode to ground at the input. Next to the fuse it looks like an overvoltage protection diode, and if so it should be a zener with a breakdown voltage of about 15-20V? But the Kingsley part looks a lot bigger than these zeners, so I wonder what the real purpose of this diode is.
Maybe somebody has a clue about these questions? Thanks!
(https://i.postimg.cc/yJV9bD0X/smps.jpg) (https://postimg.cc/yJV9bD0X)
Quote from: Rob Strand on June 02, 2024, 08:04:04 PMIt's not obvious to me what the cause is. I can only speculate. If you build a large number of boards and they all fail then it could be the layout or the diode batch or a design issue. If a different batch or supplier of diodes fixes it the it's the diode, otherwise perhaps it's the layout or design.
When you have one-off failure it could be anything. If you replace the diode with different diodes it might look like it's the diode type but it could be the diode and replacing it with the same diode type would also work.
Early on you mentioned it was getting hot. I'm assuming it's diode that's getting hot. That would be a significant piece of information. When you convert low voltage to high voltage with a boost converter the duty is very low and requires a high current pulse. It's possible the inductance value is too small. It could even be getting near saturation are bumping the current up.
Yes it could be a lot of reasons why it might have blown, from poor design to bad parts. First heard of the diode blowing from this video.
https://youtu.be/qB8FpGTyVTw?t=755
Think this may be the schematic.
(https://forum.retrotechnique.org/uploads/default/original/3X/0/c/0c7b774eb2a0ea487e1a32d52d86040ddc43db05.png)
And just for kicks.
(https://cee-gee-gee.com/YH11068/P1010303B.jpg)
Some information here.
https://www.diyaudio.com/community/threads/high-voltage-boost-converter-sacrilege.310526/page-4
There is an FCC requirement that any piece of equipment that is not battery-operated must pass conducted emissions testing at a cost of thousands of dollars unless all generated frequencies are below 9000 Hz. Someone on this forum said he made quiet power supplies that operated below 9000 Hz but did not inject any power supply noise into the circuit. This might be worth it if you intend to produce pedals for sale. I have forgotten who it was but it was in a post from this year. I would be interested in finding out what circuit topology he was using so I could determine if it was suitable for specific purposes.
FSFX (https://www.diystompboxes.com/smfforum/index.php?action=profile;u=59307) - he used to bait antonis.
https://www.diystompboxes.com/smfforum/index.php?topic=130799.0
https://www.diystompboxes.com/smfforum/index.php?topic=130687.0
https://www.diystompboxes.com/smfforum/index.php?topic=108238.0
I thought I had a 12V 5A laptop supply to use but it seems to be limiting the current. Would never have tried it but at 9.5V in I was still getting 337V out at 16W. I have a few other supplies but none over 2.7A with 12V. I did get 30W out of the module with a fan on, may have to stop by a thrift store to see if I can find a higher current one.
(https://i.imgur.com/4SjS2Wb.jpg)
Quote from: printer2 on June 03, 2024, 12:27:19 PMYes it could be a lot of reasons why it might have blown, from poor design to bad parts. First heard of the diode blowing from this video.
https://youtu.be/qB8FpGTyVTw?t=755 (https://youtu.be/qB8FpGTyVTw?t=755)
Think this may be the schematic.
Unfortunately the devil is in the details of with these type of issues. If we look at what we know:
- the diode fails
- the diode gets hot
- the output current is relatively low compared to the diode forward current rating.
- the diode failed at the high voltage setting in the video.
(Unfortunately the high voltage setting is also when the output current
is highest in the video.)
I'm thinking the problem is related to diode reverse-recovery.
The design itself uses a transformer, unlike the some of the transformerless designs in the thread. That usually helps reduce stresses in the parts.
As to why the US3MC fails and the UF4007 does not. That's not so obvious.
Something else about the video. It looks like the converter was set to a voltage
where the output no longer increases. This is like the switchmode equivalent of
regulator dropout. Some designs can stress the parts under those conditions as
some parts get driven too hard trying to keep the output voltage regulated (which
it can't do). It's best to choose adjustable pots so that can't happen.
Quote from: Rob Strand on June 04, 2024, 08:14:51 PMQuote from: printer2 on June 03, 2024, 12:27:19 PMYes it could be a lot of reasons why it might have blown, from poor design to bad parts. First heard of the diode blowing from this video.
https://youtu.be/qB8FpGTyVTw?t=755 (https://youtu.be/qB8FpGTyVTw?t=755)
Think this may be the schematic.
Unfortunately the devil is in the details of with these type of issues. If we look at what we know:
- the diode fails
- the diode gets hot
- the output current is relatively low compared to the diode forward current rating.
- the diode failed at the high voltage setting in the video.
(Unfortunately the high voltage setting is also when the output current
is highest in the video.)
I'm thinking the problem is related to diode reverse-recovery.
The design itself uses a transformer, unlike the some of the transformerless designs in the thread. That usually helps reduce stresses in the parts.
As to why the US3MC fails and the UF4007 does not. That's not so obvious.
Something else about the video. It looks like the converter was set to a voltage
where the output no longer increases. This is like the switchmode equivalent of
regulator dropout. Some designs can stress the parts under those conditions as
some parts get driven too hard trying to keep the output voltage regulated (which
it can't do). It's best to choose adjustable pots so that can't happen.
I am a toddler when it comes to SMPS's, I can follow an explanation of how the circuit works but have only read about a couple of topologies and that was a number of years ago. That said, the diode reverse recovery was something I thought might be an issue. The UF4007 has the same spec. as the US3MC, is the UF4007 much faster than it is listed at or is the US3MC slower than it should be? Or is it something else? I don't know. I did realize it was my supply that was shutting down due to overcurrent rather than the module.
Just a thought. The bipolar version of the module has an extra diode and capacitor both running half wave. Do you think the diode will have a better chance of survival if I make the output circuit a full wave rectifier? With two diodes passing current to the capacitor both would be getting half the current. While the total heat dissipated is the same for the same board area the spot temperature at the diode should be lower. Unless there is something I am missing.
Quote from: printer2 on June 05, 2024, 02:38:14 PMI am a toddler when it comes to SMPS's, I can follow an explanation of how the circuit works but have only read about a couple of topologies and that was a number of years ago. That said, the diode reverse recovery was something I thought might be an issue.
The fact the diode overheats with load currents much less than the forward rating is a key piece of info to blame reverse recovery. I probably should know the answer but I don't play with SMPS's every day so it often takes me half a day to reset my mind.
QuoteThe UF4007 has the same spec. as the US3MC, is the UF4007 much faster than it is listed at or is the US3MC slower than it should be? Or is it something else? I don't know.
Not all diodes are alike. The recovery process is complicated and the reverse recovery figure combines at least two effects. Those two effects might be different for each diode. Given the converter is right on the point of diode failure a factor of two difference in the details could save it!
QuoteI did realize it was my supply that was shutting down due to overcurrent rather than the module.
When your PSU current limits it will lower the output voltage from the set value. That could set-up a situation similar to setting the SMPS output voltage to something it cannot meet - a similar condition to what happens in the video except you are reducing the input instead of increasing the output.
QuoteJust a thought. The bipolar version of the module has an extra diode and capacitor both running half wave. Do you think the diode will have a better chance of survival if I make the output circuit a full wave rectifier? With two diodes passing current to the capacitor both would be getting half the current. While the total heat dissipated is the same for the same board area the spot temperature at the diode should be lower. Unless there is something I am missing.
The way a boost converter works is power is transferred on only one of the switch cycles. It has to work that way for either polarity output. The way you get a negative polarity output is to flip the connection of the winding (The diode doesn't actually get flipped. It usually gets moved to the other side of the winding).
I have a feeling the extra diode winding + diode + cap would do little. With such a small dummy load on the negative polarity output you would just get a little more loss on the negative output diode but the positive side would pretty much do what it did before - perhaps an insignificant fraction less.
A general trend with switching devices is throwing bigger parts at a problem doesn't always fix the problem. You can imagine someone building an SMPS, frying the diode, then updating the design with a high current diode. In reality it can actually make the true problem worse. Higher rating parts are slower. If the problem is related to switching losses the losses will get worse but perhaps the larger package diode handles the losses and gives the appearance of fixing the problem. A better solution might be to stick with a smaller rating diode which is faster.
Quote from: Rob Strand on June 05, 2024, 08:57:02 PMQuoteJust a thought. The bipolar version of the module has an extra diode and capacitor both running half wave. Do you think the diode will have a better chance of survival if I make the output circuit a full wave rectifier? With two diodes passing current to the capacitor both would be getting half the current. While the total heat dissipated is the same for the same board area the spot temperature at the diode should be lower. Unless there is something I am missing.
The way a boost converter works is power is transferred on only one of the switch cycles. It has to work that way for either polarity output. The way you get a negative polarity output is to flip the connection of the winding (The diode doesn't actually get flipped. It usually gets moved to the other side of the winding).
I have a feeling the extra diode winding + diode + cap would do little. With such a small dummy load on the negative polarity output you would just get a little more loss on the negative output diode but the positive side would pretty much do what it did before - perhaps an insignificant fraction less.
A general trend with switching devices is throwing bigger parts at a problem doesn't always fix the problem. You can imagine someone building an SMPS, frying the diode, then updating the design with a high current diode. In reality it can actually make the true problem worse. Higher rating parts are slower. If the problem is related to switching losses the losses will get worse but perhaps the larger package diode handles losses and gives the appearance of fixing the problem. A better solution might be to stick with a smaller rating diode which is faster.
I took a look at the schematic after reading your explanation and it seems obvious. I looked through Digikey's selection and not a lot of fast high voltage diodes that will fit. Found the following, physically small but doable.
https://www.digikey.ca/en/products/detail/vishay-general-semiconductor-diodes-division/VS-E7FX0212-M3-I/22145760
Quote from: printer2 on June 06, 2024, 04:10:39 PMI took a look at the schematic after reading your explanation and it seems obvious. I looked through Digikey's selection and not a lot of fast high voltage diodes that will fit. Found the following, physically small but doable.
https://www.digikey.ca/en/products/detail/vishay-general-semiconductor-diodes-division/VS-E7FX0212-M3-I/22145760 (https://www.digikey.ca/en/products/detail/vishay-general-semiconductor-diodes-division/VS-E7FX0212-M3-I/22145760)
You did well just the same. Modern diode technology and it certainly looks like it's made for the job. You can die from information overload selecting parts these days!
Quote from: Rob Strand on June 06, 2024, 05:09:56 PMYou did well just the same. Modern diode technology and it certainly looks like it's made for the job. You can die from information overload selecting parts these days!
Took half a day trying to find one that ticks off all the boxes. And now that I am ordering, what else should I pick up? So much for a quick use of someone else's technology.
Quote from: printer2 on June 07, 2024, 10:56:23 AMTook half a day trying to find one that ticks off all the boxes. And now that I am ordering, what else should I pick up? So much for a quick use of someone else's technology.
Don't feel alone, it's a common occurrence. Sometimes you get find a good part but it's 35wks lead time. You can see why companies stick to using the same chips/parts.
Quote from: Rob Strand on July 14, 2023, 08:53:06 PMInteresting the timing cap value is 390pF. The trace I had for the through-hole version quoted 100pF to 270pF
which seemed a bit low. The 390pF (say 330pF to 470pF) seems more likely.
The SMD board has an SMD inductor. From what I can work out it's 12mm x 12mm x 8mm high. Wurth sometimes have different inductor heights for the same footprint and the inductors have different specs.
From what I can work out it's a,
Wurth (WE), PD SMT Shielded Power Inductor, 744770222
Isat 1.49A
RDC 247m ohm typ., 390m ohm max
Here's the through-hole trace from the web with a few fixes,
(https://i.postimg.cc/CRjqGhrh/kingsley-HV-converter-through-hole-sch-V11.png) (https://postimg.cc/CRjqGhrh)
There's no heatsink on the MOSFET.
Any idea why the current sensing resistor was omitted?
Could it be added to replace the fuse? Not as a part of the reverse-polarity protection of course, only to protect the circuit in case of a short.
Would a current sensing resistor change noise level? Probably not, I guess.
Regards,
Markus
Quote from: markusw on June 10, 2024, 06:22:06 AMAny idea why the current sensing resistor was omitted?
Could it be added to replace the fuse? Not as a part of the reverse-polarity protection of course, only to protect the circuit in case of a short.
Would a current sensing resistor change noise level? Probably not, I guess.
One question I can answer myself :icon_redface:
The current sensing resistor won't help in case of a short since there is a path to gnd via the diode. ;)
The other questions remain.
Markus
Quote from: markusw on June 10, 2024, 12:58:50 PMOne question I can answer myself :icon_redface:
The current sensing resistor won't help in case of a short since there is a path to gnd via the diode. ;)
The other questions remain.
That's pretty much it!
Quote from: markusw on June 10, 2024, 06:22:06 AMAny idea why the current sensing resistor was omitted?
Could it be added to replace the fuse? Not as a part of the reverse-polarity protection of course, only to protect the circuit in case of a short.
Would a current sensing resistor change noise level? Probably not, I guess.
The current limit on these devices only limits the switch peak current; the app note sets the limit to the built-in switch current limit. I don't think it does a good job of limiting the output current and won't protect against short circuits. So it's more parts, lower efficiency due to more voltage drop across the sense resistor, then ultimately not full protection.
Quote from: Rob Strand on June 10, 2024, 11:03:37 PMThe current limit on these devices only limits the switch peak current; the app note sets the limit to the built-in switch current limit.
Thank you very much for your help!
As I understand the current limiting resistor is calculated via the saturation current of the inductor. Therefore, it should prevent saturation of the inductor. Wouldn't this improve the design?
Not sure what you mean with "the app note sets the limit to the built-in switch current limit." :icon_redface:
Quote from: Rob Strand on June 10, 2024, 11:03:37 PMSo it's more parts, lower efficiency due to more voltage drop across the sense resistor, then ultimately not full protection.
I did a LTSpice simulation of the Kingsley SPMS with 320V output, 130k load (~2.5mA) and a 0.3 ohm current sensing resistor (limiting peak current to ~1A). Dissipation of the current sensing resistor in average is only roughly 15 mW versus 0.8W of the load.
Quote from: markusw on June 11, 2024, 01:25:22 AMAs I understand the current limiting resistor is calculated via the saturation current of the inductor. Therefore, it should prevent saturation of the inductor. Wouldn't this improve the design?
Not sure what you mean with "the app note sets the limit to the built-in switch current limit." :icon_redface:
It limits both the inductor current and the switch current. You can limit to whatever breaks first, or lower currents if you want to impose a different limit. The MC34063 application note (and some other parts like the TL497) just highlight the maximum current is set by the switch (build into the part).
Under normal conditions the boost converter limits it's own current by design (the peak inductor current is set by the on-time and the inductor value). The idea is not to rely on the current limit to set the operating peak current.
Quote from: markusw on June 11, 2024, 01:25:22 AMI did a LTSpice simulation of the Kingsley SPMS with 320V output, 130k load (~2.5mA) and a 0.3 ohm current sensing resistor (limiting peak current to ~1A). Dissipation of the current sensing resistor in average is only roughly 15 mW versus 0.8W of the load.
It seems quite low but believable. I haven't checked by any hand calculations.
I guess from a design perspective you have to ask what are you protecting against. It becomes a bit of a value judgement.
Things like design issues should be fixed before relying on protection, eg. the diode fault in this thread. In normal operation for a tube there's not a lot to go wrong and short the supply rails. In other applications the parts in the load circuit can fail and short and then wreak havoc on the supply. For cases like that I have been very grateful designers have put in supply protection because it limits the damage and helps debug the fault. In other cases the supply or load has failed and shorted out a whole lot of parts. In one case I had to fix the fault and rewind the inductor then make sure I got it all right before power up. That's where you wish some form of safety valve was put in place to minimize the damage.
Quote from: Rob Strand on June 13, 2024, 07:08:17 PMIt limits both the inductor current and the switch current. You can limit to whatever breaks first, or lower currents if you want to impose a different limit. The MC34063 application note (and some other parts like the TL497) just highlight the maximum current is set by the switch (build into the part).
Under normal conditions the boost converter limits it's own current by design (the peak inductor current is set by the on-time and the inductor value). The idea is not to rely on the current limit to set the operating peak current.
Again, thank you very much for your help!
Without the current sensing resistor and a 330p timing cap the LTSpice sims predict slightly above 0.8 A peak current (roughly the same as with the 0.3 ohm sensing resistor), with a 470p cap peaks are around 1.2A without the current sensing resistor (all with 320V output voltage and 130k load).
Without the current sensing resistor there is a ~3A peak at start up, that is prevented by current sensing resistor.
Don't know whether this only a LTSpice artifact.
Quote from: markusw on June 15, 2024, 05:15:53 AMWithout the current sensing resistor and a 330p timing cap the LTSpice sims predict slightly above 0.8 A peak current (roughly the same as with the 0.3 ohm sensing resistor), with a 470p cap peaks are around 1.2A without the current sensing resistor (all with 320V output voltage and 130k load).
Without the current sensing resistor there is a ~3A peak at start up, that is prevented by current sensing resistor.
Don't know whether this only a LTSpice artifact
You could be on to something there. The fact the current limit helps reduce the start-up current is hint it is real. Power up surges are a source of components failing. Silicon power devices will handle somewhat higher than rated currents for short periods. It's when those currents are too high it can cause failures in the field, not always today but some time in the future. A lot of modern devices implement soft starting. (I've got some simulations but they are on another computer.)
Something you can try on spice is change the power source delayed pulse. Then you can see what happens at power up. You can also experiment with the risetime of the pulse to see how it affects the start-up current. That gives an idea what is real and what the cause is.
In a real circuit you might have a wall-wart with 20 ohm output impedance feeding a 1000uF cap when the main is switched on there is a natural soft start. However, if you have 2A a switch mode power supply wall-wart the output impedance is lower and the start-up isn't as soft. Yet another scenario is the wall-wart is already powered-up then you plug it into the DC converter. The different power scenarios can affect what currents flow at power up.
For the boost converter you can get a start-up current pulse from the inductor through the diode to the output cap. That current pulse cannot be fixed with with the current limit (other than the current limit resistor adding some series resistance and limiting crazy high currents. Normally the inductor DC resistance will be higher.)
Quote from: Rob Strand on June 15, 2024, 08:19:17 PMYou could be on to something there. The fact the current limit helps reduce the start-up current is hint it is real. Power up surges are a source of components failing. Silicon power devices will handle somewhat higher than rated currents for short periods. It's when those currents are too high it can cause failures in the field, not always today but some time in the future. A lot of modern devices implement soft starting. (I've got some simulations but they are on another computer.)
Something you can try on spice is change the power source delayed pulse. Then you can see what happens at power up. You can also experiment with the risetime of the pulse to see how it affects the start-up current. That gives you can give you an ideal what is real an what the cause is.
Thanks!
You mean delaying the start of the supply voltage and/or slowly ramping it up?
For a given set of conditions - forward current, reverse voltage, and turnoff time - each reverse recovery generates the same amount of heat. The average heat produced in a part depends on how many times you do a reverse recovery per second. That is, raw switching frequency matters. The more often you do a reverse recovery, the hotter your switching things get.
I learned this while in the process of causing a neat, round hole to be melted through the top of a steel cased Motorola TO3 switching transistor. :-(
The higher the switching frequency, the lower the recovery losses per switching has to be to keep the heat down to the same level.
Quote from: markusw on June 16, 2024, 11:30:46 AMYou mean delaying the start of the supply voltage and/or slowly ramping it up?
Yes for the examples I gave. However, in power electronics you will also come across deliberate schemes where the PWM is limited on startup which lowers start-up current, reduces glitches and of course starts-up slower.
Quote from: R.G. on June 16, 2024, 03:24:53 PMFor a given set of conditions - forward current, reverse voltage, and turnoff time - each reverse recovery generates the same amount of heat. The average heat produced in a part depends on how many times you do a reverse recovery per second. That is, raw switching frequency matters. The more often you do a reverse recovery, the hotter your switching things get.
For the frying diode issue: As output voltage increases the boost converter frequency rises so the losses go up as per your comments. In the video that mechanism is compounded by the increased output voltage and current. All three mechanisms are increasing and adding to the stress on the diode.
Quote from: Rob Strand on June 17, 2024, 06:29:34 PMFor the frying diode issue: As output voltage increases the boost converter frequency rises so the losses go up as per your comments. In the video that mechanism is compounded by the increased output voltage and current. All three mechanisms are increasing and adding to the stress on the diode.
Yes, that is true.
I've done a small foray into the recovery losses of the two kinds of diodes in play. Honestly, my thumbnailing of the recovery losses from the data sheets don't turn up losses that ought to cause hot diodes. My best guess right now is that there is something odd going on to run up the recovery losses on the diodes.
Not sure what that is right now, but I can't yet reconcile the kinds of recovery losses needed for the heat described with the circuit conditions, even given the higher voltage and current. Really, the currents and voltages are not that high. I'm relating this to the voltages and currents on snubber circuits on the primaries of flyback power converters that produce tens of watts. Something isn't correlating yet.
In my ongoing quest for numbers, I found a couple of references to refresh me:
https://www.google.com/url?sa=t&source=web&rct=j&opi=89978449&url=https://www.st.com/resource/en/application_note/an5028-calculation-of-turnoff-power-losses-generated-by-a-ultrafast-diode-stmicroelectronics.pdf&ved=2ahUKEwjL3dLz6eOGAxV7lokEHcj5ADEQFnoECBYQAQ&usg=AOvVaw1MEkPifG4MtC-bTcceJyCw
and
https://www.vishay.com/docs/98280/howtocalculatepowerlossesingen5diodes.pdf
as well as a couple of others.
When I get a little free time I'll try to thumbnail the losses to see if the described voltage, current and frequency cause an honest heat rise for the diodes, or if there is obviously something funny.
This one is quite good.
https://www.infineon.com/dgdl/an-989.pdf?fileId=5546d462533600a40153559fa625124d
It covers everything and has some numbers. The diode data is more complete than most diodes. In the literature you will find reverse recovery is discussed in terms of switch loss and not so much the diode loss but that document covers both.
Quote from: R.G. on June 17, 2024, 09:13:39 PMWhen I get a little free time I'll try to thumbnail the losses to see if the described voltage, current and frequency cause an honest heat rise for the diodes, or if there is obviously something funny.
Agreed. I plugged in some numbers back when this issue came up and I wasn't convinced; need to find time to go over it in detail. The symptoms are quite convincing through. The problem with reverse recovery is it depends on the circuit. The values in the datasheet are more values from test cases, not hard numbers to plug into formulas for all circuits.
Something which did seem odd. Suppose we fry the diode due to overheating at 150degC. The thermal resistance of the diode was something like 45 degC/W. So we need 3.3W of diode loss. I'm not seeing that type of number in the calculations.
Another point was the long tracks on the PCB could have enough inductance to raise the reverse voltage (which is in the doc I posted). [Actually at the time of the thread I didn't realize there was a transformer until the schematic was posted. The transformer leakage inductance will contribute.]
Quote from: Rob Strand on June 17, 2024, 10:41:25 PMSomething which did seem odd. Suppose we fry the diode due to overheating at 150degC. The thermal resistance of the diode was something like 45 degC/W. So we need 3.3W of diode loss. I'm not seeing that type of number in the calculations.
That's one of the biggies, all right. The whole circuit doesn't seem to be moving enough power to make the diodes overheat, especially tie to wide, flat PCB areas. I've toasted my fair share of fast - and slow! - diodes, but this situation seems to not carry enough sheer power in bad conditions to overheat the diodes. Maybe. But it sure seems not to.
When that happens, I get out my textbooks, rules of thumb, pyramid amulets and slide rules and start looking at numbers.
Quote from: R.G. on June 17, 2024, 11:46:30 PMThat's one of the biggies, all right. The whole circuit doesn't seem to be moving enough power to make the diodes overheat, especially tie to wide, flat PCB areas. I've toasted my fair share of fast - and slow! - diodes, but this situation seems to not carry enough sheer power in bad conditions to overheat the diodes. Maybe. But it sure seems not to.
I can imagine some extra reverse voltage due to the transformer output going negative during the on cycle and some ring from the transformer leakage over-voltaging the diode. That would just zap the diode. But for the diode to get progressively hotter as the output voltage and load is dialed up that has to be something else. Very low output currents so forward loss unlikely too.
Back again. Going to try putting lipstick on a pig. I spent hours looking for a suitable replacement heatsink and finally gave up and searched through my junk. Cut apart a cpu heatsink and managed to mount it in a fraction of the time. Ordering replacement diodes as I do not trust the ones blowing. For good measure thinking of stuffing the distance between the transformer and the heatsink with compound to wick off heat from the transformer.
(https://i.imgur.com/Pi5EKTz.jpg)
Quote from: printer2 on June 23, 2024, 04:13:52 PMFor good measure thinking of stuffing the distance between the transformer and the heatsink with compound to wick off heat from the transformer.
It's not worth the mess. Ferrite is a poor conductor of heat, and so is heatsink compound. So in the end it's not going to achieve much cooling the entirity of the transformer. Besides transformers can take a bit more than you might expect. A better idea would be to put the edges with the heatsink and transformer in a more open air in the enclosure, away from any corners.
Quote from: Rob Strand on June 23, 2024, 05:56:12 PMQuote from: printer2 on June 23, 2024, 04:13:52 PMFor good measure thinking of stuffing the distance between the transformer and the heatsink with compound to wick off heat from the transformer.
It's not worth the mess. Ferrite is a poor conductor of heat, and so is heatsink compound. So in the end it's not going to achieve much cooling the entirity of the transformer. Besides transformers can take a bit more than you might expect. A better idea would be to put the edges with the heatsink and transformer in a more open air in the enclosure, away from any corners.
It does not take much arm-winging to get me not to smear that stuff, hate it. I was not thinking of the heat bothering the transformer but rather to reduce any heat that gets to the board and therefore the diode. Did not think it would be a magic bullet but every quarter ounce eventually adds up to a pound (used to build and crash RC gliders). I am hoping to have the module on the bottom of a Tweed styled chassis with a heat shield between it and the tubes. Also being on the outside I want to use some perforated metal to shield against radiated EMI.
I did some quick (and very dirty) estimation on the switching diode losses for the circuit in question. The US3M looks like it ought to work, mostly, which might account for the stories of it working until the voltage gets high. The UF4007 surprised me in that it has a shorter trr than the US3M, 75nS vs 85nS; I had expected the purpose-packaged US3M to be better, but both are pretty good.
Both have similar thermal characteristics. The US3M has a thermal resistance of 26C/W to its terminal when mounted on 0.53" X 0.73" pads of 2-ounce copper. This reflects the fact that most of the heat is conducted out the terminals into the PCB copper, not through the body. The UF4007 has a similar situation, in that it specifies thermal resistance of 30C/W to its leads. It too relies on the PCB copper for heat sinking. The UF4007 doesn't specify a PCB pad or area size, only a max power dissipation of 2W.
Some of the very dirty stuff is in estimating (not calculating or simulating) the reverse recovery losses. If the diode is conducting current into a cap full of 300Vdc, then tries to stop conducting, it takes trr seconds to stop conducting. During that time, it has to conduct the reverse current and (to a first approximation) the DC voltage from the output capacitor. This is a big assumption based on thinking that the transformer capacitances and reflected voltage conditions can pull the cathode nearly to ground(ish) to get the capacitor voltage across the diode.
Another dirty assumption was that the reverse recovery current was a square pulse of 1A. The datasheets show a curve for reverse recovery current peaking at 1A, and trr being at recovery to 1/4 of that. Rather than do the integration graphically or with math in my head, I reasoned that it has to be better than 1A for trr. So each turn off event could be about 300V * 1A = 300W. This power only lasts for trr, so the energy is 300W*85nS, or 25.5uJ. Doing that at a switching frequency of 100kHz gives 2.55W. This goes down with switching frequency, of course. If it's 50kHz, the dissipation is only 1.28W, etc.
Two-ish watts in a diode with 30C/W thermal resistance would make the junction about 60C over ambient, and so yes, it gets hot. The same reasoning applies to both diodes. That makes me feel better about my intuition that there wasn't enough power being converted to heat the diodes. There is, it's just that there's quite a bit going into the diodes.
I then wondered - why does the UF4007 survive? Probably because there are two of them and they are close-enough matched to share dissipation. Or that the UF4007 semiconductor process and packaging is somehow better inside. The PCB area isn't particularly great compared to the datasheet notes on the US3M, so maybe that's an issue.
The high voltage output is an issue if my thinking isn't way off. The diode reverse losses would go up with output voltage. They would also go up linearly with switching frequency. Yeah - if my thumbnailing isn't too far off, the diode could get into heat stress and not live long because of thermal effects as the output voltage goes up. Turning the output voltage down lets it cool off. There might be an option to lower the switching frequency if the inductor can take the longer on-time, or if the inductor can be subbed for one with enough energy storage at the lower frequency.
Quote from: R.G. on June 24, 2024, 11:23:23 AMI did some quick (and very dirty) estimation on the switching diode losses for the circuit in question. The US3M looks like it ought to work, mostly, which might account for the stories of it working until the voltage gets high. The UF4007 surprised me in that it has a shorter trr than the US3M, 75nS vs 85nS; I had expected the purpose-packaged US3M to be better, but both are pretty good.
Both have similar thermal characteristics. The US3M has a thermal resistance of 26C/W to its terminal when mounted on 0.53" X 0.73" pads of 2-ounce copper. This reflects the fact that most of the heat is conducted out the terminals into the PCB copper, not through the body. The UF4007 has a similar situation, in that it specifies thermal resistance of 30C/W to its leads. It too relies on the PCB copper for heat sinking. The UF4007 doesn't specify a PCB pad or area size, only a max power dissipation of 2W.
Some of the very dirty stuff is in estimating (not calculating or simulating) the reverse recovery losses. If the diode is conducting current into a cap full of 300Vdc, then tries to stop conducting, it takes trr seconds to stop conducting. During that time, it has to conduct the reverse current and (to a first approximation) the DC voltage from the output capacitor. This is a big assumption based on thinking that the transformer capacitances and reflected voltage conditions can pull the cathode nearly to ground(ish) to get the capacitor voltage across the diode.
Another dirty assumption was that the reverse recovery current was a square pulse of 1A. The datasheets show a curve for reverse recovery current peaking at 1A, and trr being at recovery to 1/4 of that. Rather than do the integration graphically or with math in my head, I reasoned that it has to be better than 1A for trr. So each turn off event could be about 300V * 1A = 300W. This power only lasts for trr, so the energy is 300W*85nS, or 25.5uJ. Doing that at a switching frequency of 100kHz gives 2.55W. This goes down with switching frequency, of course. If it's 50kHz, the dissipation is only 1.28W, etc.
Two-ish watts in a diode with 30C/W thermal resistance would make the junction about 60C over ambient, and so yes, it gets hot. The same reasoning applies to both diodes. That makes me feel better about my intuition that there wasn't enough power being converted to heat the diodes. There is, it's just that there's quite a bit going into the diodes.
I then wondered - why does the UF4007 survive? Probably because there are two of them and they are close-enough matched to share dissipation. Or that the UF4007 semiconductor process and packaging is somehow better inside. The PCB area isn't particularly great compared to the datasheet notes on the US3M, so maybe that's an issue.
The high voltage output is an issue if my thinking isn't way off. The diode reverse losses would go up with output voltage. They would also go up linearly with switching frequency. Yeah - if my thumbnailing isn't too far off, the diode could get into heat stress and not live long because of thermal effects as the output voltage goes up. Turning the output voltage down lets it cool off. There might be an option to lower the switching frequency if the inductor can take the longer on-time, or if the inductor can be subbed for one with enough energy storage at the lower frequency.
So you are saying, "It depends?"
Just kidding. Your back of a napkin analysis is at the level I can absorb easily. Now I have a a few more data-points to use with Digikey's selection.
Thank you.
"Oh, you were looking for Elegance? Keep walking and it will be three doors to the left. This is Just Get The Damn Thing Done"
Only got it up to 19W with the power supply I have at the moment and the 350V caps I got in the circuit. The brick does not like when the module is set for 350V on startup and shuts down and restarts. Turning down the set voltage and then turning it up it does not get upset. Set it to 350V and 50mA through a 6V6 in SE.
The heatsinks are pretty cool. With ambient being 24 C I have 37 C on the foil (0.2 mm, 0.1 mm that I folded on itself and run some solder between them) and 33 C on the big old heatsink after an hour and a half. OK that was fun, how about if I put the original heatsink back? This time 19.7W out I got 35.2 C on the foil, 38 C on the heatsink. Seems the foil really does help. I will have to run more power through it, maybe even with a higher supply voltage.
(https://i.imgur.com/Sz9Q6Fn.jpg)
Swapped the 6V6 for a 6N3C Russian tube, on 310V I had 80 mA, 25W out of the module and the original heatsink heated up to 55 C. I did not get to measure the foil as the diode popped. By the time I got the thermistor on the foil it was also measuring 55 C so I am sure it got hotter.
The module with a 6V6 in SE (359V 55 mA) seems like it would be fine with my little makeshift heatsink. I need to look at replacement parts for the diode with sucking the heat away in mind.
Quote from: printer2 on June 24, 2024, 10:05:12 PM6V6 in SE (359V 55 mA)
That's a lot of heat for a 6V6. 19.7W, maybe 17.7W if you allow for bias and screen. Rated 12-14W. Also 315V. Yes, there is a report of Fender pushing modern Champs to that power zone.
Your Russian 6L6-alikes may be fine.
Quote from: PRR on June 24, 2024, 10:51:54 PMQuote from: printer2 on June 24, 2024, 10:05:12 PM6V6 in SE (359V 55 mA)
That's a lot of heat for a 6V6. 19.7W, maybe 17.7W if you allow for bias and screen. Rated 12-14W. Also 315V. Yes, there is a report of Fender pushing modern Champs to that power zone.
Your Russian 6L6-alikes may be fine.
Yeah I know, but I did not have a better way of dissipating the power. Nothing glowed red so I was not too worried. I have more tubes than I will ever need so not a big concern. About 25 NOS 12V6's in the upper left, a hand full of used also beside them.
(https://i.imgur.com/t9LNevY.jpg?1)
(https://i.imgur.com/0XjS7BV.jpg)
Quote from: R.G. on June 24, 2024, 11:23:23 AMSome of the very dirty stuff is in estimating (not calculating or simulating) the reverse recovery losses. If the diode is conducting current into a cap full of 300Vdc, then tries to stop conducting, it takes trr seconds to stop conducting. During that time, it has to conduct the reverse current and (to a first approximation) the DC voltage from the output capacitor. This is a big assumption based on thinking that the transformer capacitances and reflected voltage conditions can pull the cathode nearly to ground(ish) to get the capacitor voltage across the diode.
Another dirty assumption was that the reverse recovery current was a square pulse of 1A. The datasheets show a curve for reverse recovery current peaking at 1A, and trr being at recovery to 1/4 of that. Rather than do the integration graphically or with math in my head, I reasoned that it has to be better than 1A for trr. So each turn off event could be about 300V * 1A = 300W. This power only lasts for trr, so the energy is 300W*85nS, or 25.5uJ. Doing that at a switching frequency of 100kHz gives 2.55W. This goes down with switching frequency, of course. If it's 50kHz, the dissipation is only 1.28W, etc.
I'm not convinced about the 1A estimate agreeing with what is happening in the circuit. The forward currents are quite low in this circuit. I was planning to get back to the cause myself but coming up some realistic estimates is actually a fair amount of effort. (I'm sure that's why you used the 1A value - I get it, we something to work with!!)
There's also a technicality in that the power loss is some factor less than the V*I product. These factors creep in when V and/or I changes with time; factors like 1/2, 1/4, 1/6. All fair enough but it does push the power dissipation estimate down.
The Trr values in the datasheets are only valid for the test conditions in the datasheet. So when the current and voltages in the circuit are different to the test circuit you get different recovery times.
This video has a good demo of the effect of the circuit on the actual currents and recovery times.
You can pretty much dial up any recovery time you want.
#201: Basics of Reverse Recovery Time in a Diode
6:00 Demo Start
7:40 Increase forward bias/Forward current ==> recovery time to increase.
8:00 Increase reverse bias ==> decrease recovery time
The flat-bottomed recovery waveforms are a result current limiting due to the signal source resistance.
The schematic seems to have a bug. I think the 100nF timing cap is incorrect. IIRC the predicted switch time for 100nF is very low, less than 10kHz. Your 50kHz to 100kHz values make more sense, that's the type of numbers I was using last week. It's not possible to resolve the switch time unless someone measures the switching waveforms or the cap.
I tried to put some basic numbers to the design. However, we can only go far on specifics since we don't know the switch frequency and we don't know the winding ratio or inductance of the transformer.
Nonetheless, I've assigned some values which allow 20W output at 200V. I chose an on-time of 10us roughly ball-parking a 50kHz to 100kHz switch frequency range. With the values shown, the duty cycle and discontinuous conduction off time are feasible numbers.
Bottom line is:
- it's possible the forward diode current is the cause of the problem.
- that would mean the transformer winding ratio isn't quite right.
For the simulation I manually change the secondary inductance and the switching period to maintain 200V output.
The results are in the text.
(https://i.postimg.cc/dhtM0stS/basic-numbers-V10-flyback-converter.png) (https://postimg.cc/dhtM0stS)
As for the video posted earlier showing the diode frying. Perhaps the simultaneous increase in output voltage and load current (due to fixed resistive load) just pushes the *forward* diode current too far.
Quote from: Rob Strand on June 26, 2024, 03:19:30 AMI'm not convinced about the 1A estimate agreeing with what is happening in the circuit. The forward currents are quite low in this circuit. I was planning to get back to the cause myself but coming up some realistic estimates is actually a fair amount of effort. (I'm sure that's why you used the 1A value - I get it, we something to work with!!)
I'm not convinced either. It's a really dirty estimate. :-)
I could rationalize it by the filter cap pouring charge back into the junction for a few nanoseconds while something mysterious was happening in the transformer or primary. Anyway, the recovery current is probably not dependent on the forward current, only that the junction won't actually support any reverse voltage until the charges in the junction are swept out by reverse current flow. The filter cap could do that for 1A, maybe; depends on what's sucking on the anode side of the diode keeping the anode more negative than the couple of hundred volts on the filter cap.
About all I can say for myself is that it does seem to come out with a result that matches the symptoms, kind of.
Edit; hit send too soon.
Yes, a forward current of an amp or two in a diode with a 1.8v max forward drop for roughly the same 85nS gets to about 3W as well. Could be a transformer issue. If I had transformer specs I could do a much less rough estimate for that. Maybe it's both forward and reverse at the same time.
About now, if I were trying to get this to work, I'd whip in a massively bigger horse. Mouser has the Vishay VS-E5TX1512S2L-M3 for $2.12 each, and it might be coaxable into the pads on the PCB, although it's not a real fit. It's rated for 15A and only 29ns trr. It would make for interesting theater.
Quote from: R.G. on June 26, 2024, 10:24:16 PMI'm not convinced either. It's a really dirty estimate. :-)
I could rationalize it by the filter cap pouring charge back into the junction for a few nanoseconds while something mysterious was happening in the transformer or primary. Anyway, the recovery current is probably not dependent on the forward current, only that the junction won't actually support any reverse voltage until the charges in the junction are swept out by reverse current flow. The filter cap could do that for 1A, maybe; depends on what's sucking on the anode side of the diode keeping the anode more negative than the couple of hundred volts on the filter cap.
About all I can say for myself is that it does seem to come out with a result that matches the symptoms, kind of.
Edit; hit send too soon.
Yes, a forward current of an amp or two in a diode with a 1.8v max forward drop for roughly the same 85nS gets to about 3W as well. Could be a transformer issue. If I had transformer specs I could do a much less rough estimate for that. Maybe it's both forward and reverse at the same time.
The symptoms looked more like reverse issues to me too.
Well I had a closer look at the details and 1A reverse current isn't unreasonable. For 300V out and 40W output load: With the 500uH secondary I could get a peak reverse current of 1.3A with 1.1W dissipation due to the reverse current. With a 200uH secondary both those figure are approximately doubled. The dissipation due to forward currents is quite low, like 250mW or so. Given we know nothing of the transformer ratio that's about as far as I'm willing to go. To get to the bottom of it someone needs to pin down the details of the transformer and the switch frequency.
FWIW, I set-up the transformer with a lower inductance so the Toff time was longer. Using half the previous values, ie. Lp=8.5uH and Ls=250uH, the reverse power in the diode dropped considerably. It highlights the need for details of the actual unit.
Finally got back to it. Frequency 72.5 kHz, 300V 79 mA. Flattened out a couple of pennies (make sure they are copper kiddies) and did some fancy soldering as well as painted them and the original heatsink flat black, still bare copper in the picture.
(https://i.imgur.com/MzTYrnT.jpg)
(https://i.imgur.com/NfeuDho.jpg)
I did not get a lot of time on it but the heatsinks felt a little warm to the touch.
Quote from: printer2 on June 29, 2024, 10:38:34 PMpennies (make sure they are copper kiddies)
Where do you get those?? US Cent has been 2.5% Copper since 1982, our VietNam era. Bad drives good out of circulation so you won't find two of the 95% Copper cents today (unless you pay as a collector). I think Canada has quit cents. UK Penny is steel with electroplate copper.
Copper roof flashing is a thing and I have seen small squares sold individually for chimney step-flashing, and have used that material as fins on LM377 chips.
Quote from: PRR on June 29, 2024, 11:17:43 PMQuote from: printer2 on June 29, 2024, 10:38:34 PMpennies (make sure they are copper kiddies)
Where do you get those?? US Cent has been 2.5% Copper since 1982, our VietNam era. Bad drives good out of circulation so you won't find two of the 95% Copper cents today (unless you pay as a collector). I think Canada has quit cents. UK Penny is steel with electroplate copper.
Copper roof flashing is a thing and I have seen small squares sold individually for chimney step-flashing, and have used that material as fins on LM377 chips.
After my copper foil experiment (scraps from shielded a MRI suite) I thought I would just buy a small piece of sheet, the smallest they had was a one foot square piece, clicked on the tab, OMG $97. Forget that. Yes we got rid of the penny up here a number of years ago, had a coffee can of pennies from back in the day. We went over from copper to plated steel in 1998 (think some might have been zinc alloy or something, a magnet would not pick them up). Pre-1982 were thicker, I separated about 50 of them when I thought that would be more than enough. I flattened them to get a little more area but afterward though just putting a flat spot on the penny and soldering it in place would do. I was concerned about the transformer pins but I would just trim the area of the penny close to to them and call it a day.
I need to do a little more testing to see how much I can pull out of the modules without too much effort. Also if they will upset the 2.7A laptop supply as they do the 2A one I was using that gets overloaded on startup. Maybe they might be worth using if you have a couple of cents to throw at them (sorry, I could not help that).
Changing/debasing the coinage is no fun, and I see Canada went through both Zinc and Iron (and a dash of Nickel) on the way. I see that throwing pennies in a jar is a long tradition, so they may not be that hard to find. (But check your copper spot-price first!)
You won't want to hear that decades ago I finished much of a kitchen in copper sheet. Even that may have been a lucky buy: I remember trimming with "coppertone" which was dyed Aluminum. (They just retired the MRI trailer here with a much bigger unit; should I have been down there the night before "helping"?)
EDIT: bad timing..... https://www.youtube.com/shorts/Lm6_xOXndfk
Quote from: printer2 on June 29, 2024, 10:38:34 PMFinally got back to it. Frequency 72.5 kHz, 300V 79 mA. Flattened out a couple of pennies (make sure they are copper kiddies) and did some fancy soldering as well as painted them and the original heatsink flat black, still bare copper in the picture.
Quote from: printer2 on June 29, 2024, 10:38:34 PMI did not get a lot of time on it but the heatsinks felt a little warm to the touch.
Very cool.
I'll try to match something with your numbers. I'm assuming oscilloscope waveform is the output ripple at the first cap? (As it sure looks like the diode current passing through the output cap ESR.)
It would be good to get the PWM output from the chip. It's not possible to extract the MOSFET on-time from the diode current waveform. The diode current waveform is when the switch is off but it is shorter than the total off time. So we don't know the on-time or the total off-time.
Almost forgot I found this,
https://dalmura.com.au/static/YH11068A.pdf
It quotes 75kHz operating frequency, which is in agreement with you measurement. I need to look at the datasheet for that chip. I was assuming it's constant off time and variable off time, which is how the MC34063 works, but it could be different.
Nonetheless I might be able to narrow the times based on the 8V to 32V input and 40V to 390V output. Your 72.5 kHz, 300V 79 mA (and 12V in?) would then pin things down in the middle of range.
Quote from: Rob Strand on June 30, 2024, 11:05:44 PMQuote from: printer2 on June 29, 2024, 10:38:34 PMFinally got back to it. Frequency 72.5 kHz, 300V 79 mA. Flattened out a couple of pennies (make sure they are copper kiddies) and did some fancy soldering as well as painted them and the original heatsink flat black, still bare copper in the picture.
Quote from: printer2 on June 29, 2024, 10:38:34 PMI did not get a lot of time on it but the heatsinks felt a little warm to the touch.
Very cool.
I'll try to match something with your numbers. I'm assuming oscilloscope waveform is the output ripple at the first cap? (As it sure looks like the diode current passing through the output cap ESR.)
It would be good to get the PWM output from the chip. It's not possible to extract the MOSFET on-time from the diode current waveform. The diode current waveform is when the switch is off but it is shorter than the total off time. So we don't know the on-time or the total off-time.
Almost forgot I found this,
https://dalmura.com.au/static/YH11068A.pdf
It quotes 75kHz operating frequency, which is in agreement with you measurement. I need to look at the datasheet for that chip. I was assuming it's constant off time and variable off time, which is how the MC34063 works, but it could be different.
Nonetheless I might be able to narrow the times based on the 8V to 32V input and 40V to 390V output. Your 72.5 kHz, 300V 79 mA (and 12V in?) would then pin things down in the middle of range.
I did not vary the input voltage (12V) or the load on the module. I did step it up from 100V in 50V steps. At 100V it had a time of 59 kHz and from 150V to 300V it was 72 kHz. I need to make a proper jig so I can clip in my probe, as it was I was only able to clip onto the copper heatsink on the output cap side (only one cap, the other is on the input).
I don't know when I can get back to this, I kind of lost three weeks of my life due to health issues and have a big backload of stuff that needs to be done. But sometimes (heck, most of the time) I get fixated on something and waste time on something rather than more important stuff so I may get to it in a timely matter yet.
Quote from: printer2 on July 01, 2024, 12:02:23 PMI did not vary the input voltage (12V) or the load on the module. I did step it up from 100V in 50V steps. At 100V it had a time of 59 kHz and from 150V to 300V it was 72 kHz. I need to make a proper jig so I can clip in my probe, as it was I was only able to clip onto the copper heatsink on the output cap side (only one cap, the other is on the input).
That might be enough to fill in the gaps. The key thing being the variable frequency. I'm assuming a resistive load of 300/79mA = 3.8k, so at 100V the load is 100/3.9k = 26mA.
And yes, it's often easy to do some quick poking around but there comes a point where you have to set-up everything like a scientific experiment - and that wasn't the idea!
QuoteI don't know when I can get back to this, I kind of lost three weeks of my life due to health issues and have a big backload of stuff that needs to be done. But sometimes (heck, most of the time) I get fixated on something and waste time on something rather than more important stuff so I may get to it in a timely matter yet.
I hear you there ... I've be out for 9 months (and no, I wasn't pregnant :icon_mrgreen: ).
At 300V as before.
(https://i.imgur.com/Reiwtz5.jpg)
(https://i.imgur.com/aRyRJFg.jpg)
(https://i.imgur.com/ufKe6GA.jpg)
OK cool, I'll give it another shot.
Quote from: Rob Strand on July 03, 2024, 12:07:54 PMOK cool, I'll give it another shot.
C1 might be 3 nF, R6 1k. I did a rough calibration of my meter's capacitance when it went out but I am ordering some 1% caps. I could not find my camera tripods so I was poking around with my right hand, had my near sight glasses on with a pair of cheaters over them (eyes getting too old for surface mount), SLR in my left hand trying to keep in focus and push the darn button. It was an interesting evening.
Quote from: printer2 on July 03, 2024, 12:37:14 PMC1 might be 3 nF, R6 1k. I did a rough calibration of my meter's capacitance when it went out but I am ordering some 1% caps. I could not find my camera tripods so I was poking around with my right hand, had my near sight glasses on with a pair of cheaters over them (eyes getting too old for surface mount), SLR in my left hand trying to keep in focus and push the darn button. It was an interesting evening
Is that the clock resistor and cap? There does seem to be a problem on with the clock capacitance. I'm seeing C8 on the schematic you posted earlier (no idea if the PCB has designators at all, or that the schematic matches them.)
https://www.diystompboxes.com/smfforum/index.php?topic=130852.msg1284850#msg1284850
For my rough simulation I stripped everything back. I force the switching frequencies to match what you measured. I don't even have the chip in the simulation. As drawn, the schematic has no voltage clamps or snubbers so I didn't add any. I do get quite a bit of ringing at some points in the circuit.
Quote from: Rob Strand on July 03, 2024, 08:21:23 PMQuote from: printer2 on July 03, 2024, 12:37:14 PMC1 might be 3 nF, R6 1k. I did a rough calibration of my meter's capacitance when it went out but I am ordering some 1% caps. I could not find my camera tripods so I was poking around with my right hand, had my near sight glasses on with a pair of cheaters over them (eyes getting too old for surface mount), SLR in my left hand trying to keep in focus and push the darn button. It was an interesting evening
Is that the clock resistor and cap? There does seem to be a problem on with the clock capacitance. I'm seeing C8 on the schematic you posted earlier (no idea if the PCB has designators at all, or that the schematic matches them.)
https://www.diystompboxes.com/smfforum/index.php?topic=130852.msg1284850#msg1284850
For my rough simulation I stripped everything back. I force the switching frequencies to match what you measured. I don't even have the chip in the simulation. As drawn, the schematic has no voltage clamps or snubbers so I didn't add any. I do get quite a bit of ringing at some points in the circuit.
Oops, wrong schematic of the same board. R25, C10. I got the 1k but might as well ignore the 3 nF value I got(schematic value 0.1 uF). I just spun a trimmer so the display on my meter matched the reading on my meter but with only one point of calibration I would not bet any money on it. Some 1% caps on order to do it properly.
(https://forum.retrotechnique.org/uploads/default/original/3X/0/c/0c7b774eb2a0ea487e1a32d52d86040ddc43db05.png)
On the ringing, I grabbed this picture earlier on yesterday but if you ask me where I could not be sure now. I think it might have been at 100V out without a load. Thought it might be due to no load, my spaghetti wiring, who knows so I forgot about it.
(https://i.imgur.com/aCSqh2h.jpg)
Since I got those pictures I decided to live dangerously and bumped up the voltage to 390V and 73 mA, 29W. Before I had a chance to scope anything, 'pop'. I felt the copper fins after and they did not seem overly warm. Puts me out of commission until I get my order in. At least now I can clean up the corner of my desk
OK, just a small update. Got my parts in, decided to go with a 6A fast diode that should be able to pass more current before heating up too much. The diodes I picked are just big enough to straddle the copper pads on the board and I thought end of story. But after getting them I saw they are not as tall as the original part and soldering on a copper heatsink to each side is problematic. I thought these could be a low cost somewhat easy way for a beginner to get into high voltage tube amps but I am starting to think maybe not. The jury is not out yet, still going to try blowing up more parts. The original part is on the right.
(https://i.imgur.com/bxSawPc.jpg)
But I think with my little penny trick I should be able to feed a SE 6V6 without it blowing. On the bench is one thing, in an amp another, and we are not quite there yet. I made a little aluminum enclosure around the module so it could be outside the chassis and have a supply of fresh air.
(https://i.imgur.com/dYkFlqN.jpg)
Here is the original on the right, the 6A surface mount one with an attempt to solder on the copper pennies after some reworking, the left a 15 A part but the pads dictate how it is soldered. Which sucks as the bolt to mount a heatsink is right over the transformer pins. Tried to figure out how to cool it and in the end I decided to CA glue a piece of aluminum on it.
(https://i.imgur.com/4FeE22k.jpg)
(https://i.imgur.com/umsU5Uc.png)
Painted the bare heatsink then decided to test it. Ran it at 350V and 80 mA for 28W. I stuck a thermister on the original heatsink and it got to 50 C with the rate of increase not really changing too much so I did not think it would stabilize to soon. The aluminum plate on the diode was also approximately the same temperature as the other one. Since I only have two of these diodes I decided to stop it and put a larger heatsink on the mosfet. Cranked it up again and the temperatures stabilized at 46 C.
(https://i.imgur.com/VtSNUkF.jpg)
It seemed happy enough now with the room temperature of 25 C. I then took the temperature of the transformer. The displays only go up to 70 C and the transformer was above that. This has me thinking that might be why the two heatsinks were roughly the same temperature. I planning on mounting the module upside down, already did above, might not be a good idea with the transformer heat rising. I am going to try the module with the stock part and my pennies for a 6V6 (actually 12V6) in single ended as a small Champ. If it works I may just leave it but for a higher power amp I will be going to the BYC15-1200PQ (TO220).
Quote from: printer2 on July 16, 2024, 06:42:39 PMI decided to stop it and put a larger heatsink on the mosfet. Cranked it up again and the temperatures stabilized at 46 C.
A bigger heatsink will be far more effective than painting etc.
Quote from: printer2 on July 16, 2024, 06:42:39 PMIt seemed happy enough now with the room temperature of 25 C. I then took the temperature of the transformer. The displays only go up to 70 C and the transformer was above that. This has me thinking that might be why the two heatsinks were roughly the same temperature. I planning on mounting the module upside down, already did above, might not be a good idea with the transformer heat rising.
Yes the hot transformer doesn't help. To me having the diode under the transformer knee-caps any attempt to remove heat from both. I like the TO-220 version but there some issues. The region between the PCB and the aluminum plate is going to be very ineffective for removing heat from the transformer or the diode. One side of the plate is essentially unused
halving doubling the thermal resistance. Pushing the diode down onto the PCB doesn't really help either. If you use a finned TO-220 heatsink it will maintain or improve the area but it will also have a smaller foot print and allow for better cooling of the TO-220 and the bottom of the PCB. Even one with shallow fins it could help things along. On idea is to mount the TO-220 diode to the main heatsink. Another is to cut some tracks and move it to the another edge of the PCB away from the Tx but then you will need another heatsink and have to mount the heatsink somehow. A kind of in between version would have the heatink going upto the edge/corner of the PCB where there is no Tx, then leave the area under the Tx open. That would mean putting the TO-220 on the edge if the heatsink which would be OK provide the heatsink material is thick enough. A larger gap under and around the PCB also helps.
My apologies for not getting back to the simulation. It's on my list of things to do. I've been trying to finish some stuff off.
Quote from: Rob Strand on July 16, 2024, 09:04:14 PMQuote from: printer2 on July 16, 2024, 06:42:39 PMI decided to stop it and put a larger heatsink on the mosfet. Cranked it up again and the temperatures stabilized at 46 C.
A bigger heatsink will be far more effective than painting etc.
Quote from: printer2 on July 16, 2024, 06:42:39 PMIt seemed happy enough now with the room temperature of 25 C. I then took the temperature of the transformer. The displays only go up to 70 C and the transformer was above that. This has me thinking that might be why the two heatsinks were roughly the same temperature. I planning on mounting the module upside down, already did above, might not be a good idea with the transformer heat rising.
Yes the hot transformer doesn't help. To me having the diode under the transformer knee-caps any attempt to remove heat from both. I like the TO-220 version but there some issues. The region between the PCB and the aluminum plate is going to be very ineffective for removing heat from the transformer or the diode. One side of the plate is essentially unused halving the thermal resistance. Pushing the diode down onto the PCB doesn't really help either. If you use a finned TO-220 heatsink it will maintain or improve the area but it will also have a smaller foot print and allow for better cooling of the TO-220 and the bottom of the PCB. Even one with shallow fins it could help things along. On idea is to mount the TO-220 diode to the main heatsink. Another is to cut some tracks and move it to the another edge of the PCB away from the Tx but then you will need another heatsink and have to mount the heatsink somehow. A kind of in between version would have the heatink going upto the edge/corner of the PCB where there is no Tx, then leave the area under the Tx open. That would mean putting the TO-220 on the edge if the heatsink which would be OK provide the heatsink material is thick enough. A larger gap under and around the PCB also helps.
My apologies for not getting back to the simulation. It's on my list of things to do. I've been trying to finish some stuff off.
No worries about the sim, think any fast diode will work if they remain cool. Until I checked the transformer temperature we thought it was the diode heating itself to be the problem. Or it could have been and now the power is high enough for the transformer to be the next issue. Earlier when I was thinking of the diodes I wondered if using two and the winding to produce the negative voltage (for the ones that give a +/- supply) to halve the power going through a single diode. Would flip the second diode and capacitor around and combine the two through a pair of resistors to hopefully balance out the current through the two.
(https://forum.retrotechnique.org/uploads/default/original/3X/0/c/0c7b774eb2a0ea487e1a32d52d86040ddc43db05.png)
Now wondering if using the second winding would decrease the transformer losses. Just depends if the heat is more from copper losses or from the core. Let's see, P = I
2 x R, drop R in half and you dissipate less power heating up the place. Looking at the schematic, seems like there might be another winding for the primary. Another place to reduce copper loss? Would they be significant?
This sure is becoming more involved than I originally anticipated. Yes the T0-220 can be mounted on a common heatsink but the existing mosfet would have to be replaced with one having longer leads to extend over the edge of the circuit board. Or use the replaced heatsink in the same place as it was but under the board and bolt the diode to it. If I could find bolts long enough and the same diameter as the ones used to hold the original heatsink to the board (some modules do not even have the heatsink mounted and just hanging off the mosfet) I could sandwich the two heatsinks together.
Oh wait, I am looking at a board with the original heatsink on it and the one I pulled. Could cut another section of the cpu heatsink I used for the bigger replacement heatsink and do the same thing. It would be convenient using the original one though, already has the hole tapped and ready to mount the TO-220 diode. Just depends if the original one is up to the task. Was the bigger heatsink just helping to get rid of the heat allowing me to draw more current? I know a normal person would just put a fan on a stock module and be done with it. I dislike fans as they can fail, replace them on speed drives at work from time to time.
edit: just a quick note, flipped the diode and capacitor around on a dual voltage out board. Weird response, almost half the voltages as compared to when it is configured as a negative supply..
Quote from: printer2 on July 17, 2024, 12:06:54 PMNow wondering if using the second winding would decrease the transformer losses. Just depends if the heat is more from copper losses or from the core. Let's see, P = I2 x R, drop R in half and you dissipate less power heating up the place. Looking at the schematic, seems like there might be another winding for the primary. Another place to reduce copper loss? Would they be significant?
Yes it's a good idea. It could make a small improvement. Not sure if it will work due to technicalities - see below. Windings in switching supplies are tricky because of the proximity effect and losses due to the gap in the core. However in this case the unused winding is already present. Unsed copper is bad, so you might as well put it to use.
QuoteThis sure is becoming more involved than I originally anticipated. Yes the T0-220 can be mounted on a common heatsink but the existing mosfet would have to be replaced with one having longer leads to extend over the edge of the circuit board. Or use the replaced heatsink in the same place as it was but under the board and bolt the diode to it. If I could find bolts long enough and the same diameter as the ones used to hold the original heatsink to the board (some modules do not even have the heatsink mounted and just hanging off the mosfet) I could sandwich the two heatsinks together.
I'm not really surprised. The unit has one or more minor design faults which come from design decisions early on - basically there is no margin at maximum load. It's always hard to patch fix these types of problems from the outside other than operate at outputs less than rated load You can do *something* and you can try your best but the final result might not be what you would end-up with if you could start from scratch.
QuoteWas the bigger heatsink just helping to get rid of the heat allowing me to draw more current? I know a normal person would just put a fan on a stock module and be done with it. I dislike fans as they can fail, replace them on speed drives at work from time to time.
More than likely. Fans help a lot as they improve virtually all aspects of heat dissipation, that includes the transformer. It's just a pain having a fan.
Quoteedit: just a quick note, flipped the diode and capacitor around on a dual voltage out board. Weird response, almost half the voltages as compared to when it is configured as a negative supply..
You can't flip just the diode. On a flyback converter the output from each winding has to happen on the same part of a switch cycle. There's no option for a push-pull type arrangement. For the negative winding, you have to flip the winding and make sure the diode is around the right way.
(https://i.postimg.cc/r0y9WW6Z/possible-rejig-of-windings.png) (https://postimg.cc/r0y9WW6Z)
[edit: for the second case you could leave the diode with it's original direction in the negative lead. That means you can use the negative diode mounting.
(https://i.postimg.cc/WtGkjzQh/possible-rejig-of-windings-case-2-without-moving-diode.png) (https://postimg.cc/WtGkjzQh)
]
The biggest unknown is if the winding on the negative supply matches that on the positive winding. If not it might not actually contribute any current. If the negative winding was multifilar wound with the positive winding then you can parallel the winding. If you use an extra diode it might help with some mismatch, not much. A third option is to create a completely separate positive rail then combine the outputs with resistors.
Yet a fourth option, which requires *much* more care, would be to put the windings in series. That means you can utilize the second winding without worrying about the windings sharing current. However it's significant change and could fry something.
Another unknown is the pin numbers on the transformer that's going to require some checking/measurements. You can go off that schematic but is it correct and is 5,6 a pair or 5,7 the pair? Then for the negative winding why do we see pins 8,9 in parallel?
The loss in the transformer is due to the copper loss and also the loss in the ferrite. Usually both are equally significant. You can can reduce ferrite losses by reducing the switch frequency (actually increase the on time). However for the same power output if you reduce the switch frequency you need more winding current and that will increase the copper loss. Similarly you you can increase the switch frequency to trade the other way. We don't know which one is worse for this design. An easy way to do a test is to measure the efficiency. Then make a small change to the frequency up/down/or both and see which way improves the efficiency. Then perhaps try a bit more in the same direction. There's a lot of checks required to be confident this doesn't cause problems. The idea here is if the supply is more efficient it will generate less heat. However, if you want to reduce the loss in a specific part that might result in poorer efficiency - it's like the other components have to suffer more to prop-up a weak one.
Quote from: Rob Strand on July 17, 2024, 07:14:25 PMQuote from: printer2 on July 17, 2024, 12:06:54 PMQuoteedit: just a quick note, flipped the diode and capacitor around on a dual voltage out board. Weird response, almost half the voltages as compared to when it is configured as a negative supply..
You can't flip just the diode. On a flyback converter the output from each winding has to happen on the same part of a switch cycle. There's no option for a push-pull type arrangement. For the negative winding, you have to flip the winding and make sure the diode is around the right way.
(https://i.postimg.cc/r0y9WW6Z/possible-rejig-of-windings.png) (https://postimg.cc/r0y9WW6Z)
[edit: for the second case you could leave the diode with it's original direction in the negative lead. That means you can use the negative diode mounting.
(https://i.postimg.cc/WtGkjzQh/possible-rejig-of-windings-case-2-without-moving-diode.png) (https://postimg.cc/WtGkjzQh)
]
The biggest unknown is if the winding on the negative supply matches that on the positive winding. If not it might not actually contribute any current. If the negative winding was multifilar wound with the positive winding then you can parallel the winding. If you use an extra diode it might help with some mismatch, not much. A third option is to create a completely separate positive rail then combine the outputs with resistors.
I am not as good as I used to be, realized I had to move the winding around and flip the diode and cap. Then went to do other stuff and came back, flipping the cap and diode with the transformer winding forgotten. I probably wander around more looking for stuff I put down, part of the reason I am trying to get some projects done and shelve the rest and concentrate on health and wellbeing.
I marked up the board to show what needs to be done, not the easiest as the transformer needs to be pulled and the ground plane around pin 5 needs to be cut away to provide enough insulation. Pin 8 and 9 are common, a cut is needed between them, pin 9 removed and pin 8 to pin 7 as a ground point. Pin 5 can go into the pin 9 hole. Then you combine the two capacitor positive terminals together with a pair of resistors going to the amplifier circuit. I think. One other thing, I noticed the single output transformers have the windings below the surface of the transformer core and the bipolar board has the windings protruding above the surface of the core. Makes sense, why pay for more going into a part than need be if it will not be used. Would it be worth going through the trouble? I will have to mull it over, maybe a winter project, this has already taken more time than I had planned on. Good to know about the shortcomings though.
(https://i.imgur.com/NLriv2J.jpg)
And I said I would be putting this away today, hah. Popped the transformer off the board.
(https://i.imgur.com/7a8JmbU.jpg)
(https://i.imgur.com/9y3Db8B.jpg)
I took some measurements, not sure how accurate my meter is reading mH. Used a voltage divider to determine the resistances, the output side matches my meter readings on ohms. Just for kicks I fed each side of the secondary with my signal generator with a sine wave and read the voltages off my scope. from 70 kHz to 150 kHz the step up ratio roughly 16:1
(https://i.imgur.com/9QNDAYi.jpg)
Quote from: printer2 on July 19, 2024, 03:36:47 PMI took some measurements, not sure how accurate my meter is reading mH. Used a voltage divider to determine the resistances, the output side matches my meter readings on ohms. Just for kicks I fed each side of the secondary with my signal generator with a sine wave and read the voltages off my scope. from 70 kHz to 150 kHz the step up ratio roughly 16:1
Very cool. The 2.1mH/2.2mH don't look off the mark. In my previous simplified simulation I think the current started to dropped off at 500uH.
The 2.75R vs 2.17R secondary resistances means the windings aren't multifilar wound. The 2.75R is wound over the 2.15R. The 2.24mH inductance vs 2.12mH inductance are just annoyingly different that it's unclear if the voltages (turns) on the two secondary winding are the same.
The 0.11mH primary might be 0.011mh = 11uH. I had 17uH to ensure enough energy was built up in the core for a 48W output.
If we take your voltage ratio of 16, then 2.1mH should translate to 2.1mH/16^2 = 8.2uH.
To me the transformer core looks like an EFD25 or EFD30 (tell-tale sign is the winding doesn't extend past the the outer ferrite wall). I suppose you could measure the dimensions to confirm.
If you look at p15 or, IR AN1024
https://www.infineon.com/dgdl/an-1024.pdf?fileId=5546d462533600a401535591115e0f6d
Table of ratings
EFD25 20 to 30W
EFD30 30 to 50W
However, one of TDK's docs gives an EFD25 a 55W flyback rating at 100kHz,
https://www.tdk-electronics.tdk.com/inf/85/ds/b82802a.pdf
The devil is in the details on this stuff which might explain the higher TDK rating. Nonetheless t looks like the EFD25/EFD30 size is in the ball-park of the 48W rating.
I haven't tried to match up the winding resistances and inductances. It's a bit tricky as we don't know the division of the winding area between the primary and secondary. It looks like the primary is a single wire but not a *lot* thicker than the secondary. Also the size of the core EFD25 vs EFD30 is going to add to the uncertainties.
I'm guessing that the inductances you measured were the open-circuit inductances of the windings. If so, did you happen to measure the leakage inductances? I usually found that the energy stored in leakage inductance was as troublesome as switching losses.
Quote from: Rob Strand on July 19, 2024, 09:54:42 PMQuote from: printer2 on July 19, 2024, 03:36:47 PMI took some measurements, not sure how accurate my meter is reading mH. Used a voltage divider to determine the resistances, the output side matches my meter readings on ohms. Just for kicks I fed each side of the secondary with my signal generator with a sine wave and read the voltages off my scope. from 70 kHz to 150 kHz the step up ratio roughly 16:1
Very cool. The 2.1mH/2.2mH don't look off the mark. In my previous simplified simulation I think the current started to dropped off at 500uH.
The 2.75R vs 2.17R secondary resistances means the windings aren't multifilar wound. The 2.75R is wound over the 2.15R. The 2.24mH inductance vs 2.12mH inductance are just annoyingly different that it's unclear if the voltages (turns) on the two secondary winding are the same.
The 0.11mH primary might be 0.011mh = 11uH. I had 17uH to ensure enough energy was built up in the core for a 48W output.
If we take your voltage ratio of 16, then 2.1mH should translate to 2.1mH/16^2 = 8.2uH.
I just checked again, the meter gave me two different readings on the primary. On the 20 mH setting it reads 0.11 mH and on the 2 mH setting 0.012 mH, on the 200 mH setting it was reading 0.2 mH so I thought the 0.11 mH reading would be the better one to report. Also the primary has two windings in parallel.
QuoteTo me the transformer core looks like an EFD25 or EFD30 (tell-tale sign is the winding doesn't extend past the the outer ferrite wall). I suppose you could measure the dimensions to confirm.
25.5 x 25.0 x 9mm, like these easy questions.
QuoteIf you look at p15 or, IR AN1024
https://www.infineon.com/dgdl/an-1024.pdf?fileId=5546d462533600a401535591115e0f6d
Table of ratings
EFD25 20 to 30W
EFD30 30 to 50W
However, one of TDK's docs gives an EFD25 a 55W flyback rating at 100kHz,
https://www.tdk-electronics.tdk.com/inf/85/ds/b82802a.pdf
The devil is in the details on this stuff which might explain the higher TDK rating. Nonetheless t looks like the EFD25/EFD30 size is in the ball-park of the 48W rating.
I haven't tried to match up the winding resistances and inductances. It's a bit tricky as we don't know the division of the winding area between the primary and secondary. It looks like the primary is a single wire but not a *lot* thicker than the secondary. Also the size of the core EFD25 vs EFD30 is going to add to the uncertainties.
Quote from: R.G. on July 20, 2024, 09:51:44 AMI'm guessing that the inductances you measured were the open-circuit inductances of the windings. If so, did you happen to measure the leakage inductances? I usually found that the energy stored in leakage inductance was as troublesome as switching losses.
Do not know how I missed your post, I'll see about it tomorrow.
One step forward, two step back kind of day. I rewired the winding so they are in parallel. Twice, second time the charm, you know the above board and below board being backwards? Not the first time, I drilled the chassis holes for mounting the other module on the wrong side, going to look more Marshall than Fender. But I got it sorted out and did a quick check and it is making the proper voltage, tomorrow I will put it under load. Oh shoot, I soldered in the transformer already? How about a single supply transformer for the leakage inductance?
(https://i.imgur.com/A3L4UA7.jpg)
(https://i.imgur.com/mPAA1ur.jpg)
Quote from: printer2 on July 20, 2024, 10:36:24 AMI just checked again, the meter gave me two different readings on the primary. On the 20 mH setting it reads 0.11 mH and on the 2 mH setting 0.012 mH, on the 200 mH setting it was reading 0.2 mH so I thought the 0.11 mH reading would be the better one to report. Also the primary has two windings in parallel.
The meter is having some issues measuring small inductances. I'd probably trust the reading on the lowest range (2mH) of 0.012mH. Once you start measuring value below 1/10 full scale you expect some loss in accuracy but at 1/100 full scale it's unlikely many of the digits are significant. 2mH/100 = 0.02mH so even the 2mH range is pushing your luck a bit.
Quote from: printer2 on July 20, 2024, 10:36:24 AM25.5 x 25.0 x 9mm, like these easy questions.
Ok cool thanks. I'll work off that.
I did some rough checks. I did simulation with full power and 390V output running at 12V and at the minimum of 10V. At 10V the inductor current was peaking at 12A or so, and the timing with an 8.2uH primary only just makes it.
The primary inductance needs to be around 8.2uH in order to get enough power going through at full load. I'm trusting the estimate of the primary inductance based on the secondary inductances you measured and the 1:16 voltage ratio, which is around 8.6uH.
Then based on the core being EFD25 gapped core and with an assumed AL factor of 160nH/t^2, the largest standard gap, I juggled the inductance measurements and estimated that at a maximum flux density of 300mT (for ferrite) the inductor would handle 15A. In short all those numbers look pretty consistent. The 12A peak of the simulation vs 15A maximum current look fine. The core is probably operating at flux density of 300mT * 12/15 = 240mT.
https://www.tdk-electronics.tdk.com/inf/80/db/fer/efd_25_13_9.pdf
The next step was to match the DC resistances and the turns on the transformer. When I did this it looks like the transformer isn't wound to fill the winding window. However when you look at the yellow tape the winding looks pretty full. I even tried to guess the wire thicknesses from the pic and I got a similar result. So not I'm not sure what's going on.
Based on the transformer rating tables I posted earlier and we might expect an EFD25 to be a little pushed at 48W and that might explain why the transformer gets hot.
The last thing I tried was to workout how hot we expect the transformer to get. This requires a lot of work. I did some simple calculations and from what I can see the transformer shouldn't get that hot. I need to check over things. When I saw the 240mT flux density I was expecting it to heat up the core, especially for 75kHz.
Quote from: printer2 on July 20, 2024, 10:13:17 PMDo not know how I missed your post, I'll see about it tomorrow.
It's tricky to measure accurately and it's a small value so your meter might have issues. I've got some estimates in my simulation. The TDK document I posted earlier has table with some leakage values for the EFD transformer (although these need to be translated for different turns).
Found a mistake in the wiring of the secondary, fixed the image above. Ran the module at 325V, 73mA, 24W for an hour and the stock heatsink was 43 C, the core was 46 C and the stock diode with my two penny heatsink soldered on it was 32.5 C. Then I turned the pot to increase the voltage to 250V and the voltage dropped to 314V. Under no load it would get up to 390V, not under load. Decided to let it cool down and try it, still no luck. I did check the voltage of the 12V supply then, down to 11.8V, never checked it before with the other module which is a shame. Not sure what the fault could be, it seems to work fine at 325V.
I measured the transformer after I did the fix with the coil going to the correct pin and got 0.006 mH on the primary, leakage 0.001 uH (the lowest reading on the meter, I was unsure of the reading on this scale before in case the zero value was out). On the secondary I got 2.6 mH and with the primary shorted, 1.95 mH. Just for giggles I turned it up to 360V and it sat there for a moment and then dropped down to 314V. This is the only dual voltage board I have might have to order another.
Running it at 275V now, getting a lot hotter now, 60 C or more. I'll look at it again tomorrow.
I will have to check at other supply voltages but with this one at 11.7V full load the module sags. I tried three different ones, two single and the dual voltage one I modified and they all sagged with a higher load. Just eyeballing two meters I get 28-30W as it starts to droop and it goes down to 20-23W. I did the tests with them cold and ran them only as long as I needed to to get my readings so the heating is taken out of the equation.
Just did a quick test with a 16.2V supply. Set for 390V it did not sag with 104 mA out, 41W. Cool. Next is to test the heating but that will take some time and I am behind with regular life right now.
The module with my two penny heatsink is doing well in a Champ type of circuit at 350V and 45 mA from 11.5V.
(https://i.imgur.com/eOKSYXw.jpg)
Adorable! Do you notice any buzz being picked up by the output transformer?
Quote from: merlinb on August 23, 2024, 04:49:07 AMAdorable! Do you notice any buzz being picked up by the output transformer?
Absolutely none. I was concerned about conducted or radiated EMI when I started this, part of the reason for making the amp. I planned to do a P-P amp with the module and started this amp to see what pitfalls I might find. Just never thought it would take this long, thought it would only take a week. I also made the tone stack switchable from the regular Fender response to almost a flat response. It is different from using a tone stack lift in that the bass and treble controls still work but with less range. Also when switching from one to the other I do not get the massive gain boost but rather the same volume but with the hole in the response filled in.
I originally had a smaller speaker in it and was going to make this a SS amp but when I tried the speaker in the cabinet it sucked. So I shoehorned in the smallest 8" speaker I had (70V PA speaker) and thought the Blackface curve will give it some bass given the small cabinet. Or make it less boxy sounding. I have only played through it for a short time but it seems to work. It can not be mistaken for a full size cabinet but it is fine as a practice amp. At first I though switching to the flat setting sounded quite middy but after a touch of EQ it sounds fine. As it is set right now I can switch between both settings and I can not bother tweaking the EQ. I did want to use a 250k pot for the treble but when I looked in the bag that was suppose to have 250k pots in it I had 500k's. Thinking right now I guess I could have used the pot for the bass control but at the time I thought using it in the treble position would make it easier to do a point to point wiring of the stack.
(https://i.imgur.com/qSV8J7H.jpg)
With the 250k pot. The 500k circuit has a 100k instead of the 82k, 22 nF instead of the 75k (68k on the schematic, did not notice the difference before creating the image), 33 instead of the 47 nF. With the same relative signal level between both settings I do not have to bother adjusting the volume control going between either.
(https://i.imgur.com/d13GGMZ.jpg)
Otherwise it is a basic Champ other than using the 10k/500pF input filter. I spaced the controls as close to each other as I could to give me more sonic options. I would like to have an Orange Squeezer on the input and a slap back delay after the volume pot. If I feel really ambitious I might stick in a reverb instead, but these are winter projects. Right now it is sitting beside my TV and it is easy to overlook. Because the amp is so quiet the only noise I hear is from the guitar I built with a single coil pickup. Now I am going to have to put a noise canceling coil in it to get it quieter.