This idea came to me after discussing the problems with back-to-back transformers in another thread:
https://www.diystompboxes.com/smfforum/index.php?topic=121201.0
It's kind of hot off the press and I haven't tested the idea on a real set-up. Nonetheless it's kind of a nifty idea. The downside is the added capacitor has quite a large value. It might be possible to shift the cap the output side of TX2 and use a smaller value. I forgot to mention that loading on TX2 might also stuff things up a bit.
Anyway, the idea is simple: add a capacitive load between the two transformers to shift the phasing of the load current on TX1 so the power dissipation of TX1 is reduced.
The plots go: no cap, cap less than optimum, optimum cap
(https://i.postimg.cc/XrFtv1CT/Back-to-Back-Transformers.png) (https://postimg.cc/XrFtv1CT)
(https://i.postimg.cc/XBpz6sW1/Ptx1-Total.png) (https://postimg.cc/XBpz6sW1)
(https://i.postimg.cc/xq4Z3VGw/Ptx1-Primary-Secondary.png) (https://postimg.cc/xq4Z3VGw)
This one shows how the output voltage of Tx2 is increased and is closer to the primary voltage of Tx1,
(https://i.postimg.cc/D83MMdD6/Back-to-Back-Output-Voltage.png) (https://postimg.cc/D83MMdD6)
Pretty neat, Rob! I was going to ask if all of this was due to phasing problems!
Neat idea, Rob.
Did you put in the winding resistances in your sims?
You're right that transformer specs rarely tell you everything you need, and that the variations in the unspecified things can be large, and that any critical design work requires measurement for one-offs and probably custom design for any production work.
Transformer specs on magnetizing current don't help, as a lot depends on how hard the designer (or his resident bean counter) is working the iron. The higher the primary voltage, the worse this gets. The number of wire turns needed to get a primary inductance high enough to drive down the magnetizing current increases linearly with input voltage, but the core winding window is fixed for a given core size, so wire resistance becomes a bigger and bigger part of the input impedance for a transformer, even with no load.
The bean-counter's hand is heavy in this calculation. Small transformers are low profit margin items, so the pressure is to make them as cheaply as possible, and that pushes for the minimum iron and copper, but also for the minimum care in lamination stacking. The tightness of lamination stacking has a big effect on the actual value of primary inductance. In constructing some transformers from the raw copper and iron components and measuring things, I've seen a four to one difference in primary inductance dependiing on how carefully the iron was jogged together and pressed into square, level contact.
Magnetizing current is nonlinear, as the primary inductance is something of a fiction. It exists, but varies a lot as the flux density gets close to saturation, where it increases a lot. The variability of primary inductance with flux density is what gives the classical waveform of magnetizing inductance. Given the bean-counter's need to use less iron, copper, and labor, there is a strong push to simply specify magnetizing current as the value for the worst probable example that ccmes off the assembly line. You're right in that any one example of the production run may be anywhere in the distribution. They also don't tell you the distribution. :icon_eek:
Things that make this worse are the power line voltage and frequency. It's worse in 240Vac/50Hz countries than in 120V/60Hz countries. At least if the designer didn't make one version for both 50 and 60Hz countries, implying that all of them have the skinny wire and many turns that the 50Hz country needed.
It's unfortunate that the required cap would probably be larger than the transformer, given today's capacitors. Back to back electros and NP electros would have a wandering value as the capacitor ages. Film would be stable, but large.
Interesting thinking.
Note that while there is a pure inductive current, in any minimum-cost transformer there is also a very non-linear current as the peaks approach iron saturation.
http://www.jocet.org/papers/69-A30009.pdf
This paper tests a huge lump of iron, but the same trend is seen in thumb-size transformers. (In practice, KVA iron is not worked this hard due to running cost of lost power, but as R.G. says a few-VA $7 lump is often designed-down to just shy of real trouble within warranty.)
The current is not a sine. The current is not proportional to voltage (Tesla). A pure capacitor can be only a very approximate "compensation".
(https://i.postimg.cc/F7MTbvYq/RS-mag-1.gif) (https://postimg.cc/F7MTbvYq)
Yeah, but some might be enough in the very low iron cases.
I'm struck by how similar to a ferroresonant transformer you could make this by adding some external "leakage" inductors before the capacitor. I'm trying to think up (1 ) whether that could be made to work at all and (2) how the devil you'd "design" for it. The second transformer would be run in nominal saturation all the time, so the iron losses would get bigger, but the reversed-primary would give you out pseudo-square waves.
Of course, that makes what started as a (as the Brits say) cheap 'n cheerful dodge even more complicated yet, even after the film cap.
Maybe a NP would be good enough. Rob - what's the sensitivity of "goodness" to cap value?
I'll just say it. If the second transformer's load is small relative to the total (usually true for heater and B+ on small bottles), and the first transformer may be economically bought 2X oversize, there will be no smoke. It will sag, but that's not normally critical.
I've actually got stuff to do ATM.
Here's some add-on results.
It does answer some of the hanging questions and issues.
1) The big cap problem.
I mentioned it might be possible to move the cap to output of TX2 this means the cap could be a much smaller value. It also means it could be a poly-cap instead of a bipolar which will significantly improve the reliability.
HV vs LV Cap
HV Cap has 2 to 3% more dissipation in TX1 than the LV Cap.
Output voltage doesn't rise unless cap is smaller than optimum.
(https://i.postimg.cc/Xr2NwfcN/Back-to-Back-Transformers-Sch-HV-Cap.png) (https://postimg.cc/Xr2NwfcN)
(https://i.postimg.cc/0bn9sY4G/Ptx1-Total-HV-Cap.png) (https://postimg.cc/0bn9sY4G)
(https://i.postimg.cc/fkjVV0Fy/Back-to-Back-Output-Voltage-HV-Cap.png) (https://postimg.cc/fkjVV0Fy)
2) How accurate does the cap value need to be.
Given the method reduces the power dissipation in TX1 by a *factor* of 8 or so, there's plenty of scope for C to vary yet still have moderate reductions in TX1's power dissipation.
The table below for the LV cap case (ie. cap between the transformers) but I don't
expect the results to change much for the HV cap case.
Effect of smaller cap on total dissipation in Tx1
C/Copt Pdiss_tx1 / Pdiss_tx1_no_cap
0.1 0.83
0.2 0.74
0.33 0.51
0.5 0.34
0.71 0.19
1.0 0.12
C/Copt is the ratio of the actual capacitor value to the optimum capacitor value. So C/Copt = 0.33 means we are using a cap about a 3rd of the optimum value.
Pdiss_tx1 / Pdiss_tx1_no_cap the power dissipation in TX1 compared to that when there is no cap present. Smaller is better.
So we can see with with a large deviation from optimum a C/Copt = 0.33 is still reducing the power in TX1 by half.
QuoteDid you put in the winding resistances in your sims?
Yes. I'm using a T model it covers a lot of real behaviours.
Also allowed for more copper in the primary for a small transformer.
Quoteand that the variations in the unspecified things can be large
Some of the specs I have trouble believing. Like I've seen maximum Imag's quoted which are equivalent to 4 times the rating of the primary. I can't image a transformer coping with that!
QuoteTransformer specs on magnetizing current don't help, as a lot depends on how hard the designer (or his resident bean counter) is working the iron. The higher the primary voltage, the worse this gets.
One thing that puts a bound on skimping is skimped transformers generate more heat and the transformer still has to comply with winding temperature rise limit. Also (IIRC) the current standards require testing to pass at 25% overvoltage.
QuoteThe tightness of lamination stacking has a big effect on the actual value of primary inductance.
Yes, that's certainly going to spread out the range of inductance values.
QuoteMagnetizing current is nonlinear, as the primary inductance is something of a fiction.
The non-linearity is a real issue in the sense that it can dramatically increase the input current; however it is bounded as mentioned before. While non-linearity does cause the inductance to vary wildly with drive level when you fix the drive voltage to the mains voltage the span is much reduced. I'd suspect with the normally gap-less cores the previous point might determine the spread.
QuoteIt's unfortunate that the required cap would probably be larger than the transformer, given today's capacitors. Back to back electros and NP electros would have a wandering value as the capacitor ages. Film would be stable, but large.
Shifting the cap from between the transformers to the output of TX2 solves this problem. See my second post. I thought it might turn out like that. A concern is an NP cap would reduce in value with age and then the current would rise up to it's old bad value. Not a good situation.
QuoteNote that while there is a pure inductive current, in any minimum-cost transformer there is also a very non-linear current as the peaks approach iron saturation.
Yes. The way I look a it is the peaks will be there when the transformer is operating normally. The increased losses due to non-linearity would have to be covered in the transformer design in order for it to work in a normal one-transformer configuration. For the uncompensated back to back case we would see twice the current primary, as we would see in the case where there was no non-linearity.
What you don't want is the added cap to cause the transformer to be driven even deeper into saturation than normal. I have a feeling HV cap case might be better here.
I can simulate non-linear magnetics. I made a point of not doing it to show the idea works in principle. The only hassle it takes a lot of work tweaking the model to match a given sample.
QuotePretty neat, Rob! I was going to ask if all of this was due to phasing problems!
Well it clearly is! I've remember noticing the problems with back-to-back transformer back when I was about 10yo. It wasn't until I wrote that last post out in the other thread I realize it could be fixed!.
Quote from: Rob Strand on October 21, 2018, 11:48:41 PM
QuotePretty neat, Rob! I was going to ask if all of this was due to phasing problems!
Well it clearly is! I've remember noticing the problems with back-to-back transformer back when I was about 10yo. It wasn't until I wrote that last post out in the other thread I realize it could be fixed!.
But in most cases one or both transformers will be driving a cap-input rectifier that will at least partially compensate for the inductive current, no?
Quote from: merlinb on October 22, 2018, 05:40:41 AM
But in most cases one or both transformers will be driving a cap-input rectifier that will at least partially compensate for the inductive current, no?
??
'Splain dis to me. I'm confused. :)
I think Merlin was thinking like me, the B+ smoothing cap current, although after a rectifier, are phase shifted anyway? But is that only true for ripple current, which isn't enough to help?
What confused me was that in a cap input rectifier, the diode(s) conduct only at the peaks of the input sine wave. So the current is a series of peaks phase aligned with the peak AC voltage. It's peaky, but largely in phase with the voltage waveform.
The introduction of a cap between the trannies would have little effect if grossly too small, and would be a heavy capacitive load reflected to the input/primary of the first transformer in parallel with the primary inductance. So too big a cap draws a lot of primary current as well, but that affects mainly the wire losses, as the higher current has to be provided by the primary.
Things get interesting around resonance. There's the issue of what inductance counts; there is the reflection of the first transformer's primary inductance, in series with the secondary side leakage, then the cap, then the primary leakage of the second transformer, then the primary inductance of the "secondary-primary" of the second transformer, in parallel with any reflected load.
Ferros are set up to use this (mess of) effects by using a very high leakage and capacitance to resonate, store a bunch of energy bonging around in the resonance, and use that to keep the secondary side iron saturating, which amounts to clipping the secondary side voltage waveform output. In this ferros are nice because they have flat-topped output voltages very conducive to capacitive filtering unless they're specially "harmonic neutralized". They're quite dependent on the resonance to keep the flywheel resonance going, so this naturally made me think of what the resonance was doing to this.
My confusion was genuine, by the way, not rhetorical. :)
I must be doing something wrong. I tinkered with modelling transformers in the simulator and couldn't get the power in the first transformer to minimize significantly. I modeled with wire resistance, leakage inductance, nonlinear core inductance; I put a resistive load on the center connection, simulating the heater of a 12AX7, and full wave rectified, filtered, and loaded the output of the second transformer to get 100+V dc, the idea being that the middle connection would run the heaters and the high voltage output would run the plates of a preamp setup.
I set up multipliers for power into and out of the first transformer and then checked out the difference in the power in and power out. For my setup, I wound up with this difference being about 0.7W in all cases with zero or lower capacitive loading on the center low voltage connection.
Subbing in caps in the middle from 10uF up to 500uF, the power difference in the first transformer didn't change much until I got to about 100uF, and from there on up, the power into the first trannie and the power difference dissipated there went up with increases in capacitive loading. That seems to be in line with the idea that capacitive loading sucks more reactive current, which causes real I2R losses in the first transformer's wiring resistance.
What am I doing wrong?
QuoteWhat confused me was that in a cap input rectifier, the diode(s) conduct only at the peaks of the input sine wave. So the current is a series of peaks phase aligned with the peak AC voltage. It's peaky, but largely in phase with the voltage waveform.
That's how I see it as well. The fundamental is largely aligned. The peakyness results in harmonics.
Quote
The introduction of a cap between the trannies would have little effect if grossly too small, and would be a heavy capacitive load reflected to the input/primary of the first transformer in parallel with the primary inductance. So too big a cap draws a lot of primary current as well, but that affects mainly the wire losses, as the higher current has to be provided by the primary.
There's definitely going be cases where the rectifier+filter+load dominates and the added cap is going to have minimal effect.
The original motivation is the transformer Tx1 has enough rating to drive the rectifier+filter+load but the magnetization current is using up a significant capacity of Tx1, perhaps to the point where Tx1 is now overloaded. Adding the cap then buys back some of the capacity of Tx1.
I haven't looked at the effect of the rectifier+filter+load. It needs to be checked for sure. A real circuit also needs to be checked. My gut feeling the cap would help but is perhaps less effective that the simple transformer + transformer case.
QuoteI must be doing something wrong.
Maybe not.
The Hard Drive with my OS on it died yesterday after about 10 year of use(!). So I'm going to be dead in the water trying testing anything out for a few days.
I'd start with a simple linear transformer model with only the winding resistances and the magnetizing inductance.
Then gradually build up the complexity of the circuit.
Start by looking at the power dissipation for the back to back no cap case; using AC analysis.
Then add the cap and recheck the power dissipation; again using AC analysis
For AC analysis, I tend to use the expression abs(I(Rp))*abs(I(Rp)) * Rp to compute the power as currents in spice can take on complex values and some expressions give different results to using magnitudes only.
After that switch to a transient analysis. You have to make sure you are not analyzing a long start-up transient.
After that start to add the rectifier+filter. First with a small load, which should not change anything, then ramp up the load.
If your magnetizing current is small to start with then it might only take a small load before the rectifier+filter+load dominates.
> I've seen maximum Imag's quoted which are equivalent to 4 times the rating of the primary. I can't image a transformer coping with that!
Show one. I speculate that it is a VERY SMALL iron, with high surface/volume ratio. And the winder is not paying my electric bill. (And I am not charged for reactive current.) If a "stupid" design passes tests and is cheaper, why not??
I am sorry for your loss.
QuoteShow one. I speculate that it is a VERY SMALL iron, with high surface/volume ratio.
The 12V 150mA one on this page is close. Variants of that transformer have been kicking around in this country for about 45 years. The physical size varied somewhat and DC resistance of the windings varied quite a bit, maybe 2.5:1 from min to max across the variations. Some were rated a bit higher, say 2.5VA. Some of the larger core variants looked more like 4VA transformers.
https://download.altronics.com.au/files/docs_273.pdf
Here's another one with a more believable magnetization current, still quite high,
https://www.jaycar.com.au/12-6v-ct-150ma-1-9va-centre-tapped-transformer-type-2851/p/MM2006
I'm pretty sure they measure in at 7.5mA but I can't remember if that was for the small or larger iron variants.
But, yes, the crazy Imag's were for the smaller transformers.
QuoteI am sorry for your loss.
Thanks. It was going to happen one day. I did a full back-up of the OS & progs about 1 yr ago. Restore isn't as smooth as I expected. I've got strong doubts about getting the old drive back to any working state.
Quote from: Rob Strand on October 22, 2018, 05:51:07 PM
That's how I see it as well. The fundamental is largely aligned. The peakyness results in harmonics.
That was what I was getting at -the harmonics all represent reactive power which I believe must be capacitive in nature, i.e. the harmonic currents lag the voltage. I'm not sure how much this counts towards cancelling magnetising current though, since this thread is concerned with weeny little transformer VA ratings where the numbers get weird compared with 'normal sized' transformers.
But I suspect I'm overthinking it. Seems to me all you're trying to do is put a cap in parallel with the magnetising inductance, selected to resonate at 50/60Hz. This can be a big cap on the low voltage side, or a small cap on the high voltage side -the latter is therefore easier since it will be a more accurate cap.* You can only use a squinty-eyed average figure for magnetising inductance since it is so non-linear, but I dare say you can buy back a couple of VA which is worth doing if you're using a 3 to 12VA transformer for T1.
*Moreover, you can measure the magnetising current more accurately on that side of the transformer, before using it backwards.
QuoteBut I suspect I'm overthinking it. Seems to me all you're trying to do is put a cap in parallel with the magnetising inductance, selected to resonate at 50/60Hz. This can be a big cap on the low voltage side, or a small cap on the high voltage side -the latter is therefore easier since it will be a more accurate cap.*
I don't think you are over thinking it. It is a valid point. It's going to come down to specifics.
The way I look at harmonics is they are orthogonal to the fundamental because of the frequency. However reactive power in the true sense is orthogonal because of the phase and the energy is store and released. When resistances are present both increase losses.
Anyway even though the current pulses are more or less aligned with the fundamental, they aren't perfectly aligned and this might make all the difference to stuffing up the resonance. When the diodes are not conducting the circuit does look like the just the L and C sort of thing. However when the diode conducts it's whole different story. If you (roughly) think about the circuit as a switched conductance filter the primary resistance of tx2 (the one at the HV rectifier) is *permanently* connected to the filter cap through a larger resistor, one which has value Rp2' = Rp2 * (T / t_on). So that could have an effect. If it does is probably means the larger LV cap versions is slightly better. The other way to "fix" that would to add series resistance to Rp2. These circuit have a light load so that won't be a big deal.
Anyway Re using the smaller cap. My second longer post covered that. Without the rectifier present it is largely the same behaviour.
There's plenty of details to consider and things that might need to be massaged to get it to work in practice. It might even fall in a heap ;D.
> The 12V 150mA one on this page is close.
I must say I have not seen that type, with the fully plastic-enclosed primary, sold loose in the US. (I may have seen them inside products.)
Your examples are both Australian. I have not seen mag-current specs on most US-market parts. Do your electrical boffins have to account for this current? (And demand the full-enclosed winding?)
The specs look weird. 1.9VA, but 8% regulation?? I would expect >20% sag in such a tiny wind.
The low sag suggests fairly fat copper. The plastic in primary space says not a full dose of copper. This suggests not so many turns, and low inductance. Which agrees with the specified magnetizing current.
Primary load current: 2.05VA/240V= 8.5mA
Max magnetizing current: 15mA
Assuming mag-current is "typical", most of the primary current is reactive. Assuming "Max" is twice typical, load and mag currents are similar.
This does suggest a 1:1 config (no load on low-volt) might indeed run hot in the first primary.
Most of our uses for this hack are for small tubes where heater power is greatly in excess of plate power. 12AX7 runs say 1mA/plate, 0.6W of B+, and 1.8W of heater. The first transformer should be selected >2.4VA. A second tranny at 0.9VA might not be a problem; but we are at the limit of what can be wound cheaply. Simplicity suggests two of same 2.4VA transformers. Indeed the heat from double magnetizing current may be more than designed.
My path is to pick the first as 2X the estimated total. Prices in this range are soft, 2X VA will not cost 2X bucks. However a buck is a buck and another quarter inch may not fit. And if there is a "close-out", the "smart" buyer will opt for two of the same of whatever is on-sale.
OTOH--- note the Altronics PDF page wants a primary fuse for 6VA to 15VA, but not on the 2VA to 4.5V parts. This suggests they are "intrinsically safe", can be shorted forever. (All have thermal fuses too, in accord with local regs, but a bad-enough *small* transformer can't hurt itself from external shorts, only internal flaws.)
For some reason my post didn't show up here. I did some sim-ing and got odd results.
The effect of leakage inductance gets huge on small transformers. It is especially bad on side-by-side wound transformers, and it seems to be the style to do that on all smaller VA transformers. Spacing side-by-side primary and secondary windings increases it even more. At some illogically large spacings, the secondary isn't coupled to the primary at all any more. The leakage between the two coils means that all the field leaks before it gets into the secondary.
I suspect that the truly tiny transformers down in the units of VA range rely on both the primary resistance and the high leakage of side by side windings to decouple the primary and secondary under shorted-secondary conditions.
Silly me. I was sim-ing a 12V/1A trannie.
Hmmm. A 12AX7 needs 12V at 300ma for the heater, and about 1-2ma per plate, max at 100-300V. So if you do a primary to 12Vac for heaters, you need 12*0.3 = 3.6VA for a heater. The high voltage end needs at least 100V at 3ma, so with rectifier RMS losses, that runs out to about 0.48W out of the second trannie. That goes up linearly with the output voltage on the second trannie, so if you're using 240V primaries, you'd get about 1W out of the second trannie.
So at a minimum, a two-transformer setup needs to supply something like 4-5VA per 12AX7. Maybe an asymmetrical setup with a 4-6VA first transformer and a dinky 2VA second transformer would work, but in these sizes, you're long past smaller VAs making the transformers cheaper. All you'd save is likely to be a little weight, and the weight saving for tiny transformers isn't that big, either.
Here's a thought: if you have to use two transformers anyway, why not make one of them be 120:120 (or 240:120) for the high voltage, and the other 120 (or 240) to 12V or 6v for the heaters?
I too just lost a post, while the forum did an endless TLS handshake. Is it raining somewhere?
> A 12AX7 needs 12V at 300ma for the heater
No. 12V @ 0.15A, or 6V @ 0.3A. 1.89 Watts per bottle.
http://www.mif.pg.gda.pl/homepages/frank/sheets/127/1/12AX7.pdf
Other than the usual 2X mis-count, your math seems fine.
> why not make one of them be 120:120
While the catalogs go up that high, it often turns out there is No Stock of the 120V end of the range. 6V/12V iron is always in deep stock.
Doh. Yes, on 12V it uses 150ma. It's when it's when it's on 6V that is uses 300ma.
I guess it's been too long since I've designed in a 12AX7. :)
Excuse my absence. Give me a couple of days to get back on track.
I've got the flu + throat infection.
Computer is broken and I am broken - life is real great atm.
My condolences. Flu is no laughing matter. Get better - the transformers will be waiting for you.
Have weak honey and vinegar tea. (Manuka Honey is costly but special.) I'll have one with you. Get better!!
Thanks for the well wishes. I'm nearly back on track. It's not a good when it takes 40 mins to eat breakfast cereal you would normally scoff down in 3 or 4 mins if you had to rush off!
Just got my original drive back up and running with a lot of effort, tricks and help from some in-depth pages on the web (I'm indebted to the guys who wrote those pages!). There was no way I was going to attempt to fix that problem with a foggy head.
Quote
> The 12V 150mA one on this page is close.
I must say I have not seen that type, with the fully plastic-enclosed primary, sold loose in the US. (I may have seen them inside products.)
Your examples are both Australian. I have not seen mag-current specs on most US-market parts. Do your electrical boffins have to account for this current? (And demand the full-enclosed winding?)
The way these things have been build over the last 45 years has varied enormously. The plastic around the primary is a recent addition. So is quoting Imag. This definitely was not quoted in the old days, not even by the more reputable brands.
These days I do see Imag being quoted here and there. RS Components were quoting Imag on a number of transformers. Other specs are indirect in that they quote power dissipation at no load and maximum efficiency.
QuoteThe specs look weird. 1.9VA, but 8% regulation?? I would expect >20% sag in such a tiny wind.
The low sag suggests fairly fat copper. The plastic in primary space says not a full dose of copper. This suggests not so many turns, and low inductance. Which agrees with the specified magnetizing current.
Yes that's exactly what is happening. Over history, not all versions had 8% reg. The trick to get better regulation is have a low turns and high Imag. It actually works well because the load current is out pf phase with Imag. Suppose the full load output reference to the primary is Is' =Imag then the drop across the primary is only Rp * sqrt(Imag^2 + Is'^2) = 1.4*Rp*Imag, so the change in the voltage drops is 0.4*Rp*Is'. If Imag was small we would see a drop of Rs * Is'. A further trick used by some era's of the transformer was to drive the iron close to saturation. That pushes-up the no load Imag. However when a load is placed across the transformer the drop across primary resistance Rp cause the depth of saturation to be reduced and that less the current shifts from Imag to Is' with lower change in the drop across Rp.
QuoteThis does suggest a 1:1 config (no load on low-volt) might indeed run hot in the first primary.
I'm not sure how hot the Altronics ones get. For the others in the past the smaller core-sizes got hot and the larger core sizes were not very hot at all.
QuoteMost of our uses for this hack are for small tubes where heater power is greatly in excess of plate power. 12AX7 runs say 1mA/plate, 0.6W of B+, and 1.8W of heater. The first transformer should be selected >2.4VA. A second tranny at 0.9VA might not be a problem; but we are at the limit of what can be wound cheaply. Simplicity suggests two of same 2.4VA transformers. Indeed the heat from double magnetizing current may be more than designed.
My path is to pick the first as 2X the estimated total. Prices in this range are soft, 2X VA will not cost 2X bucks. However a buck is a buck and another quarter inch may not fit. And if there is a "close-out", the "smart" buyer will opt for two of the same of whatever is on-sale.
IMHO, connecting one TX to another back to back is pot luck. You don't know how much free capacity is left because Imag of the second transformer can use-up or exceed the capacity of the first one. That's what inspired the capacitor solution. It's a means of gaining enough capacity back so a small load can be placed on the second transformer.
QuoteThis suggests they are "intrinsically safe", can be shorted forever. (All have thermal fuses too, in accord with local regs, but a bad-enough *small* transformer can't hurt itself from external shorts, only internal flaws.)
I think they rely entirely on thermal fuses and cannot be shorted forever (eventually they will overheat an blow the thermal fuse). There's be a movement to use thermal fuses for small Tx's. However, yes, there are some which are designed for indefinite shorts. They tend to be the ones with poor regulation. IIRC the 1.9VA units don't quite make it; not unless you drop the ambient temperature.
FWIW, I did some back of the envelope calculations regarding the small 2VA transformers. Based on the typical DC resistances found on these transformers it seems if we try to optimize the number of turns in order to produce the least power dissipation from the transformer the magnetizing current comes out around 10mA to 12mA or so for 2VA. It depends a lot on the iron but at least these numbers point to some sort of optimum close to what we see in practice. I'm still stretching to believe the high value Imags from Altronics though.
Just to get ball rolling: To the output side of Tx2 I added a bridge rectifier, 10uF cap and approx 1mA load using the circuit with the HV cap.
I then plotted the total dissipated power from a transient analysis:
Click to Enlarge:
(https://i.postimg.cc/phQ6cjHv/Trans-Ptx1-Total-HV-Cap-Rect-10u-F-approx-1m-A-DC-Load.png) (https://postimg.cc/phQ6cjHv)
From these results it shows:
- pretty clear the added cap does in fact reduce the power dissipation on Tx1 quite a bit.
- the added cap required for minimum power dissipation in Tx1 is quite close to cap required when Tx2 is unloaded.
- changing the value confirms we are close the optimum for minimum power dissipation in Tx1
So at this point I'd conclude the idea of adding the cap does in fact work when a rectifier load +filter is present.
QuoteThe effect of leakage inductance gets huge on small transformers. It is especially bad on side-by-side wound transformers, and it seems to be the style to do that on all smaller VA transformers. Spacing side-by-side primary and secondary windings increases it even more. At some illogically large spacings, the secondary isn't coupled to the primary at all any more. The leakage between the two coils means that all the field leaks before it gets into the secondary.
I'll have to have a look at that.
Guys,
I have a problem, my preamp is working wonderfully well, am I required to post the results of some measurements? ??? ;D
I've added a cap at the high voltage side of the second transformer of my preamp but unfortunately didn't gain much from it. When adding a 100nF cap the voltage went up from 238 Volts to 240 Volts. But it might as well be the change of voltage from the supply line over time.
Then I did some measurements.
Both of my transformers are Amplimo ringcores, the first one is a 2x12V, the second one is a 2x15V.
The single ECC83 in the preamp was disconnected from high voltage but filaments and other s.s. circuitry on -/+ 12V were still connected. The secondary of T2 was still connected to the rectifier (4 s.s. diodes,cap) + r.c. filter (10k,44uF) + resistor for slow discharge (1M1).
Voltage
Primary T2: 28V AC
Secondary T2: 182V AC
Rectified high voltage: 245V
That's lower than I expected, 182 *sqrt(2) -2 = 255 or 230 x12/15 *sqrt(2) -2 = 258.
Current
But the thing I don't understand is the AC current through primary T2: Too low to be bothered! Some inconsistent readouts way below 1mA in 20mA and 200mA position of the multimeter. Actually there's a more substantial flow of current through primary T2 right after the preamp is switched on but my digital multimeter doesn't allow me to figure out how much peak current exactly.
Will the voltage up transformer (T2) cause a higher peak current to occur?
Is there a relation between lower rectified voltage than expected and low current through primary T2?
Some other reasoning to get some peace of mind...
Neither one of the transformers seems to heat up when the preamp is on. I don't get a sensation of power dissipation when I touch the transformers.
I've measured the 2x12V when I purchased it, it was 28 volts unloaded at the secondary.
The datasheet for T1 mentions: 2x 12V/ 1.25A and "All voltages are under full ohmic load". I interpret this information
that in case the 24V secondary was loaded by a 19.2 ohms resistor 1.25A would flow and the voltage across the secondary should most likely have dropped from 28V to 24V. As shown, the primary of T2/ secondary of T1 measured 28V AC, not a sign of a stressed out T1.
cheers,
PP
QuoteI've added a cap at the high voltage side of the second transformer of my preamp but unfortunately didn't gain much from it. When adding a 100nF cap the voltage went up from 238 Volts to 240 Volts. But it might as well be the change of voltage from the supply line over time.
The aim is to reduce the input current and stop Tx1 overheating, not to increase the output voltage.
The cap value needs to be tuned to suit your transformer. The optimum value could be vastly different to what I have posted. I only posted an example. (For the sake of others, a 120V transformer equivalent to my example would required a cap size will be 4 times without considering different transformers.)
A simple way to tune it would be to measure the input current then adjust the cap size until the minimum current is found. This requires mains measurements.
QuoteBut the thing I don't understand is the AC current through primary T2: Too low to be bothered! Some inconsistent readouts way below 1mA in 20mA and 200mA position of the multimeter. Actually there's a more substantial flow of current through primary T2 right after the preamp is switched on but my digital multimeter doesn't allow me to figure out how much peak current exactly.
Maybe measure the DC current on the HV side first to get an idea what load current you are dealing with. The rms AC current is typically just under two times the DC current.
QuoteNeither one of the transformers seems to heat up when the preamp is on. I don't get a sensation of power dissipation when I touch the transformers.
That's a good sign but you often have to wait 1hr or more for things to heat-up.
QuoteI've measured the 2x12V when I purchased it, it was 28 volts unloaded at the secondary.
That's a fairly typical result.
QuoteThe datasheet for T1 mentions: 2x 12V/ 1.25A and "All voltages are under full ohmic load". I interpret this information
that in case the 24V secondary was loaded by a 19.2 ohms resistor 1.25A would flow and the voltage across the secondary should most likely have dropped from 28V to 24V.
Yes transformer ratings apply to resistive load. Normally the voltage is higher than the nominal voltage with no load then drops to the rated voltage when the maximum resistive load is applied. Some don't quite work this way.
QuoteAs shown, the primary of T2/ secondary of T1 measured 28V AC, not a sign of a stressed out T1.
Given you measured the no load voltage then yes it looks like it's not dropping much and being overloaded.
--------------
Edit:
It just occurred to me your transformer is 30VA. When you get to larger transformers the back-to-back loading issue becomes less of a problem because the magnetizing current to the rated current ratio becomes smaller.
Is this the transformer?
https://www.amplimo.nl/images/downloads/ds%20standardrange/17012.pdf
(or maybe
https://www.amplimo.nl/images/downloads/ds%20standardrange/18612.pdf
)
I found this data on toroid transformers, see page 7
http://www.nuvotem.com/en/products/pdf/Trafo%20Catalog%20-%20English%20Feb-08.pdf
In the table look at the "No Load" current (presumably for 230V configuration) which is effectively the magnetizing current. 15VA: Imag = 2mA, 30VA, Imag = 2.8mA. These are extremely low in comparison to the 2VA EI core transformer I posted with Imag = 15mA.
I found these as well, see series 62000 (not 70000 as the file name implies) on p24 (p25 of PDF). Unfortunately no longer stocked by digikey,
https://media.digikey.com/pdf/Data%20Sheets/Amveco%20PDFs/70000_Series_Cat.pdf
They are small toroids and all have very low no-load current (Imag).
I did another test with a rectifier and load. This time I increase the output current to around the maximum the transformer can take (6.5mA on the DC output).
Click to Enlarge:
(https://i.postimg.cc/jL3mKpdm/Trans-Ptx1-Total-HV-Cap-Rect-10u-F-approx-6-5m-A-DC-Load.png) (https://postimg.cc/jL3mKpdm)
This test shows the cap value that minimizes the power dissipation in Tx1 is smaller than that for the unloaded (and lightly loaded) cases. The conclusion is: as you increase the load the rectifier and filter does have an effect and it does affect the resonant tuning. However all we need to do is use a smaller cap. The power dissipation in Tx1 is reduced with the added cap.
Antek tests their stuff and the criteria are published.
1500VA Toroidal Transformers
Open Circuit Test (core loss test):
TEST CONDITION: Apply variable voltage to primary coils (in parallel). Set voltages 120 and 140VAC at 60Hz. No load on secondary coils. Measure the primary current and input power.
Voltage input -- 120V 140V
Current input -- .09A .12A
Power lost ----- 12W 17W
http://www.antekinc.com/an-15445-1500va-445v-transformer/
http://www.antekinc.com/content/AN-15445.pdf
400VA Toroidal Transformers
Voltage input -- 120V 140V
Current input -- .04A .09A
Power lost ----- 4W 7W
http://www.antekinc.com/an-4tk400-400va-400v-transformer/
http://www.antekinc.com/content/AN-4TK400.pdf
25VA Toroidal Transformers
Voltage input -- 120V 140V
Current input -- .01A .01A
Power lost ----- 0.5W 0.5W
http://www.antekinc.com/an-0215-25va-15v-transformer/
http://www.antekinc.com/content/AN-0215.pdf
(no no-load data on the 10VA part)
Note that 60:1 range of rated VA is only 9:1 range of no-load current.
Quote25VA Toroidal Transformers
Voltage input -- 120V 140V
Current input -- .01A .01A
Power lost ----- 0.5W 0.5W
Effectively the same values as the Amveco (25VA, 230V, 5mA => equivalent to 10mA @ 115V).
I remember looking at ratings, resistances, temp rise, and weights of toroids in the past and they do tend to fall into a much tighter box, in terms of parameters, than EI cores.
Quote
Note that 60:1 range of rated VA is only 9:1 range of no-load current.
Yes, it's not proportional. Small Tx's always have relatively higher Imag.
Quote from: Perfboard Patcher on October 31, 2018, 07:38:32 AM
Secondary T2: 182V AC
Rectified high voltage: 245V
That's lower than I expected, 182 *sqrt(2) -2 = 255 or 230 x12/15 *sqrt(2) -2 = 258.
That's normal. The textbooks don't bother to tell you that sqrt(2) only works with
no load. Under load you need to use a figure of about 1.35 (or even less for low voltage, high current power supplies).
http://valvewizard.co.uk/psu.html
(http://valvewizard.co.uk/psu2.jpg)
Quote from: merlinb on November 01, 2018, 04:18:16 AMThat's normal. The textbooks don't bother to tell you that sqrt(2) only works with no load. Under load you need to use a figure of about 1.35 (or even less for low voltage, high current power supplies).
Schade said (drew) it best.
http://www.madbeanpedals.com/projects/_folders/Tube/schematics/UberTuber.gif
While the diodes look like tubes, they are "Ideal". (This was drawn before wide use of crystal power rectifiers.)
On this chart, 'Em' is the Peak voltage, so the 1.414 is already in there.
He varies the capacitor and the series resistor.
For almost all audio work, we will be in the right side of the capacitor scale. Note that above some minimum, capacitor value has little effect.
There is always some series resistance. Never real "small" because that would cost a lot (big iron+copper).
Taking a wild-stab: typical small iron regulation of 10% implies Rs around 0.1 of R. Note that this tops out around 0.78 of Em, so Vdc will be near 1.1 times Vrms.
In fact we have to design cap-input supplies for Idc significantly less than Iac. Assuming half current, Rs/R may be 0.05, factor Edc/Em about 0.84, maybe Vdc of 1.2 times Vac.
Even this is disappointing, we often have to use a larger transformer with lower losses.
Quote from: Rob Strand on October 31, 2018, 04:29:00 PM
Is this the transformer?
https://www.amplimo.nl/images/downloads/ds%20standardrange/17012.pdf
(or maybe
https://www.amplimo.nl/images/downloads/ds%20standardrange/18612.pdf
)
No, Rob, they're not the type numbers you mentioned, although they're practically the same transformer:
18012-2x12V-30VA
and also
08013-2x15V-15VA
Quote from: Rob Strand on October 31, 2018, 04:29:00 PM
Maybe measure the DC current on the HV side first to get an idea what load current you are dealing with. The rms AC current is typically just under two times the DC current.
I wasn't trying to measure the current (peak or stationary) per se I was looking out for too much current. I was more considering the possibility that using a back-to-back transformer could cause a high peak current when switching the preamp on. But it might as well be that the only thing that happens is the loading of the e-caps.
The aim is to reduce the input current and stop Tx1 overheating, not to increase the output voltage.
Was more some wishfull thinking from my side to squeeze the last drops of voltage out of it. Didn't know the current was already that low. See also the next quote.
Quote from: merlinb on November 01, 2018, 04:18:16 AM
Quote from: Perfboard Patcher on October 31, 2018, 07:38:32 AM
Secondary T2: 182V AC
Rectified high voltage: 245V
That's lower than I expected, 182 *sqrt(2) -2 = 255 or 230 x12/15 *sqrt(2) -2 = 258.
That's normal. The textbooks don't bother to tell you that sqrt(2) only works with no load. Under load you need to use a figure of about 1.35 (or even less for low voltage, high current power supplies).
There's hardly any load present. But at your service. I've removed the 1M1 resistor and did some new measurements. Voltage across the secondary is 184V AC and rectified voltage is 248V. Calculating with 184V gives 258V. Where's my 10 volts?
QuoteNo, Rob, they're not the type numbers you mentioned, although they're practically the same transformer
OK thanks.
QuoteWas more some wishfull thinking from my side to squeeze the last drops of voltage out of it. Didn't know the current was already that low. See also the next quote.
If you don't need to tune the power, and your transformers don't need it, then you certainly can tweak the cap to get the last drops outs. Nothing wrong with that idea.
QuoteThere's hardly any load present. But at your service. I've removed the 1M1 resistor and did some new measurements. Voltage across the secondary is 184V AC and rectified voltage is 248V. Calculating with 184V gives 258V. Where's my 10 volts?
I suspected that was the case but I wasn't sure. Both AC and DC measurements are under no load, yeah? I know I've tried to track stuff like that down before. There's an assumption when you go from AC to DC that the AC is purely sinusoidal (which relates rms to peak) which might not be the case with light loads. IIRC when you measure the AC and DC you put a tiny resistive load like 1/50th (maybe 1/10th) the maximum load and things start lining up.
For kicks I took the Jaycar Imag data and tried to work out what VA transformer survives back to back connection *without* the added cap. The basis for that is the rated output current referred to the primary (ILp= VA / Vmain) must be greater than the magnetization current Imag. In other words, the ratio ILp / Imag must be greater than 1. It actually needs to be greater than 1.1 to 1.2 to survive *without any load on the TX2* and if you want to put a load on TX2 then an even larger ratio is required.
Click to Enlarge:
(https://i.postimg.cc/qtwbpMKs/data-ILoadp-to-Imag-ratio.png) (https://postimg.cc/qtwbpMKs)
Assuming the Jaycar EI transformer data is correct we need at least an 8VA to 10VA for a back to back connection to work. The toroids from the Amveco data have high ratios for even small transformers and are clearly in the good zone. The Altronics Imag data implies we need even higher VAs.
Because of variations in transformer designs it's hard to extrapolate the results to other EI core transformers. Nonetheless the 6VA on the original McTube is dancing around the bad side of 8VA so we might expect some 6VA transformers to work and some to fail.
-------------------------------
EDIT:
Here is the graph for same series of Jaycar transformer but from the 2008 catalog.
It looks like the transformers were revised to comply with newer editions of local standards around 2008;
the specs might be for the old version?
The point where the ratio exceeds 1 is a bit higher at around 10VA; mainly from the "noisy" data. Notice the higher Imag part at 18VA. (I could not compare the dimensions of the 2008 transformers as they aren't given.)
(https://i.postimg.cc/t1hdhhHd/data-ILoadp-to-Imag-ratio-JAYCAR-EI-2008.png) (https://postimg.cc/t1hdhhHd)
I'm not going to add much to this thread until I pull some old transformers out of hibernation and measure some stuff.
Quote from: merlinb on November 01, 2018, 04:18:16 AM
That's normal. The textbooks don't bother to tell you that sqrt(2) only works with no load. Under load you need to use a figure of about 1.35 (or even less for low voltage, high current power supplies).
Maybe. Textbooks are remarkably good at telling you a lot of things, including both ideal-world, unreachable perfection and ever better approximations to the function of the real world. If they didn't do that, people would throw them away and write new textbooks that were better at explaining things. At least engineering textbooks are that way. Don't know about texts in the School or Arts and Flowers. :icon_lol:
One thing that the textbooks do tell you in some detail is that you can do remarkably accurate modelling with the T form of the transformer model. The factor of 1.414... is the peak-to-RMS number for an undistorted sine wave. Distort the sine wave, that no longer applies. This is vastly more relevant in today's world where most people have computers and when circuit modelling can be downloaded for free. It removes most of the need to understand the math from circuit design. You don't even need to understand why the square root of two is important to get the sim to match the reality to a close degree as long as you can follow some simple models. Easy availability of sim software makes the need for rules of thumb much less.
Capacitor input filters distort the sine wave because they only conduct for very limited times near the peak of the sine wave. selective voltage losses in leakage inductance and winding resistance let the effective voltage sag because of the current pulses, and this happens only during the diode conduction interval near the peak. So the combination of pulse current and series impedance with the diodes effectively flattens the peaks of sine waves. In cases where the leakage inductance and wiring resistance of transformers is small compared to the load, there is little distortion of the sine wave fed to the rectifier/capacitor filter, and the non-ideal losses come from the forward resistance of the diode and the reactive charging of the cap.
In tiny transformers run at low frequencies, you just can't wind them so they simultaneiously have negligible resistance, negligible leakage, and are cheap. So the imperfections get to being bigger and bigger parts of the reality.
Back 18BC (Before Computers, the 1970s) I had to design tranformer-rectifier-filter circuits by hand with equations, pencil, and paper. And graphs, and rules of thumb, and the grizzled veterans telling me I'd done it wrong again. :icon_lol: The output voltage from this kind of setup varies a lot - not least with the value of the load resistor and capacitor, ala Schade. I shudder to think of doing that now. Took forever. I can flip in some parts in a simulator, get really, really close to what the real result will be, and the differences are largely how accurately I measured the real-world parts for inclusion in the sim circuit.
Rules of thumb are great, if all you have are fingers and thumbs. :icon_lol:
It is interesting that the Schade charts remained the foundation of design, even re-printed in TI's 1977 regulator handbook; also Motorola 1982 (http://bitsavers.trailing-edge.com/components/motorola/_dataBooks/1982_Linear_Switchmode_Voltage_Regulator_Handbook.pdf).
There are manageable (slide-rule size) approximations which give results hardly better than rules of thumb. Today we can *try* to guesstimate all the parasitics and force SPICE to chomp it in sliced-time. (There is also PSUD which uses unique techniques, often gets real dang close, but I have seen it go off the tracks.)
"1.414" is like you take a $50,000 job. You don't ever get $50k. New suits, new house, taxi fare, after-work drinking, IRS, all take a slice.
The one I always liked was when the green newbies forgot about diode drops not being 0.7V when you're full wave rectifying 20A for the bulk supply for a 5V logic supply, and the diodes were operated well up into their resistive region. Then I could usually get them again for not remembering that a FWB always has two diode drops subtracted from the raw input voltage.
It's always a death of a thousand cuts. Ripple voltage is always down from the peak, never straddles the DC average output line. Capacitor ESR and ESL subtract disproportionately from the incoming charging pulses than from the outgoing DC average currents.
I just never want to catch my capacitors drinking with their friends.
:)
Actually, I just had another thought, not about transformer/rectifier/filter circuits, but people.
Back when I was extruded through the calculations for output voltage of a rectifier circuit, it was already so old-hat and boring that the EE student body tried strenuously to find a way not to study it. So did I. But there were few EEs in the populace, and only a small fraction of them remembered transformer/rectifier/filter stuff one second longer than turning in the final.
I find it dramatic that today, circuits and communications interests have grown so much that lay people voluntarily wonder about this, to the point that they ask for explanations. That is in my book, a remarkable success story for the rise of science in the popular mind.
Now I'm just going to sit here and count all my other blessings.
:)
> EEs ...only a small fraction of them remembered transformer/rectifier/filter stuff one second longer than turning in the final.
As you know, when they went out in the world, 99% of them never designed a rectifier. There was one guy in the division did all that all day long. (Actually he must have had other duties; even Acopian does not do that many from-scratch rectifiers.)
I consider it a fundamental skill for any electronic fiddler. Even though there is never an "exact" answer, you should have a good concept of how much windage to provide.
Quote> EEs ...only a small fraction of them remembered transformer/rectifier/filter stuff one second longer than turning in the final.
At university the ripple is always calculated using half the mains period but in reality it's probably 60% to 70% of that.
QuoteI consider it a fundamental skill for any electronic fiddler. Even though there is never an "exact" answer, you should have a good concept of how much windage to provide.
Back in the 1990's I derived some formulas which give somewhat better results. IIRC they gave results for peak and RMS currents, conduction times, regulation, ripple, allowance for transformer impedances. Took several pages to derive but the final formulas were relatively short.
I'm with RG I much prefer to plug stuff into spice. I suppose you could call it virtual fiddling.
Quote from: Rob Strand on November 03, 2018, 07:20:24 PM
At university the ripple is always calculated using half the mains period but in reality it's probably 60% to 70% of that.
For a full-wave, the run-down part of ripple is just what's left after the caps are charged, and it's asymptotic to the full half-cycle. That's only approached at nearly no load. You can pessimistically approximate ripple as dV = Idt/C where you make dt be the half cycle. The bigger the capacitor compared to the load resistance, the smaller the ripple gets. But the charging pulses can't be infinitely tall and infinitely short. From my own scoping rectifiers in the lab on +9V and +15V supplies (for regulating to 5.000V and 12.000V) I came to the conclusion that I could use 8mS out of the 8.666mS of the 60Hz half cycle, and I do, and it seems to work OK in most cases. However, my "most cases" are mostly not fed by high impedances referred to the secondary side of a chain of two or more transformers, so I don't have good intuition on the two-tiny-transformers-in-series and I always try to sim these when I build one.
The reality can be that the charge pulse time can be as little as 5% of the half cycle under loading with very large capacitances and diodes with small internal resistance. That leaves 95% of the cycle for run-down, and maybe the school thought that using 100% was conservative. Well, I guess it is.
QuoteFrom my own scoping rectifiers in the lab on +9V and +15V supplies (for regulating to 5.000V and 12.000V) I came to the conclusion that I could use 8mS out of the 8.666mS of the 60Hz half cycle, and I do, and it seems to work OK in most cases.
It does come down to specifics. Like how much ripple and the voltage of the DC rail, transformer resistances.
QuoteThe bigger the capacitor compared to the load resistance, the smaller the ripple gets. But the charging pulses can't be infinitely tall and infinitely short.
One thing I noticed is the charge pulses don't get as high as you expect because narrowness doesn't get as narrow as you would expect. When you are near the peak the voltage drop,Vin - Vcap, can't get enough charge into the cap because the current is limited by the transformer resistance I =(Vin - Vcap)/Rt. So what happens is Vcap sits a bit lower than what you expect. That does two things: the lower Vcap allows more current to flow, however, the important thing is it increases the T_on time of the rectifier somewhat. When you are near the peak it doesn't take much lowing of Vcap to take a wider slice off the top of the sine wave and T_on quickly increases. This also explains why you can have big caps and low ripple yet the power rating of the transformer doesn't need to be much different to the small cap case. If the current pulses were *really* narrow the RMS current would sky rocket and the transformer would overheat but we don't see that. This effect showed up in my old calculations. I did verify the results in pspice.
I remember thinking at the time when you understand the fine details of something the technology is probably obsolete.
> It is interesting that the Schade charts remained the foundation of design,
> There are manageable (slide-rule size) approximations
FWIW, FYI, here is Schade's paper. Not the best scan, and only 10% of a 6MB download.
https://www.americanradiohistory.com/Archive-Bookshelf/Technology/RCA/RCA-Electron-Tubes-II.pdf
(Also in Proc I.R.E. July 1943, if you have IEEE access.)
> high impedances referred to the secondary side of a chain of two or more transformers
You can get guidance from Schade's Fig 4, which covers source resistance to 100% of load resistance. That gives VDC of 35% of peak AC, so awful saggy.
Of mild interest to empty-heads: Fig 8 has V/I curves for vacuum rectifiers. Plain everyday home and broadcast bottles. At first they look "linear", but it is actually a 3/2 power law (10X over makes 31X up). We "know" this from Simplified Theory, but the deviations in apparently-measured data are smaller than the pen-line. There should be a "soft distortion" in here. (Merlin and Fender have done so, but generally below the 1V 1mA origin of this graph.)
QuoteFWIW, FYI, here is Schade's paper. Not the best scan, and only 10% of a 6MB download.
https://www.americanradiohistory.com/Archive-Bookshelf/Technology/RCA/RCA-Electron-Tubes-II.pdf
(Also in Proc I.R.E. July 1943, if you have IEEE access.)
You can get a nice copy of the original using google.
QuoteYou can get guidance from Schade's Fig 4, which covers source resistance to 100% of load resistance. That gives VDC of 35% of peak AC, so awful saggy.
There's a few things about that paper that need to be cleared-up:
- For a bridge rectifier we would use the full-wave Schade's curves. Rs remains unchanged.
- The ripple in Schade's paper is RMS not peak or the more common peak-to-peak. (Peak to peak is about 3.2 x Rms.)
- E_ac_max (~E in the paper ) is the maximum AC value with no load. Technically we should also subtract the diode drops.
I find that fig 4 doesn't line up with pspice; pspice produces a lower Eav/Eac_max value.I just check it again and it looks OK; graph is about 1% high. I must have screw up reading the graph somehow. I'm pretty sure I checked some points against pspice in the past and they were a bit off.
The ripple from fig 7 seems OK.
(Eg. 24V 1A transformer 10% reg + 5000uF cap + bridge rectifier.)
Quote from: PRR on November 10, 2018, 04:04:19 PM
(Merlin and Fender have done so, but generally below the 1V 1mA origin of this graph.)
For posterity:
(https://i.postimg.cc/F1rYGNxW/Rectifier-Characteristics.png) (https://postimg.cc/F1rYGNxW)
QuoteFor posterity:
Every time I look at those curves I think diode + resistor.
And you'd be right: posterity would appreciate knowing that you can generate your own charts by sticking an ideal diode and a resistor into the circuit and mapping the V-I curve and varying the resistor.
The posterity chart makes a great tool for fitting a semiconductor diode and resistor to the tube's action. This is, I suspect, what Weber did to make their copper top rectifier "tubes". You can get remarkably, remarkably close to a tube rectifier by "dirtying up" a semiconductor diode.
(http://)
I've seen people DIY this type of thing but I hadn't come across that product.
There's a nice pic of the innards a the bottom of this page:
http://www.diytube.com/phpBB2/viewtopic.php?f=14&t=4921
(I don't know if they other using fast diodes to reduce glitches. I doubt they add any caps across the diodes.)
QuoteYou can get remarkably, remarkably close to a tube rectifier by "dirtying up" a semiconductor diode.
I agree but I can hear those hard-core tubers chanting in the distance "get rid of that silicon".
> I think diode + resistor.
Over a w-i-d-e range, the curve of a space-charge-limited diode will deviate significantly from a semiconductor diode.
As we may only need a decade of action, and the ear is not so precise, it may be moot to us.
QuoteOver a w-i-d-e range, the curve of a space-charge-limited diode will deviate significantly from a semiconductor diode.
I suppose the advantage is it has a natural short circuit protection but I agree in the normal range the issue is probably moot.
I remember the old selenium rectifiers and copper oxide rectifiers. They were much friendlier when you get short circuits and had a tendency to limit short term overloads. I remember replacing a selenium rectifier with a silicon rectifier in car battery charger once. People often short the battery charger terminals together to check it's working. Need I say the smoke got out of the large silicon rectifier in a short space of time. The next version used a resistor but best of all was to use a high wattage bulb so the resistance increases under shorts. I noticed some commercial chargers used the light bulb as the selenium rectifiers and copper oxide rectifiers faded out. Later on they used solid state circuits and fold-back current limiting.